Propagation path estimation method and apparatus

ABSTRACT

In estimating propagation path in an OFDM receiver in an OFDM communication system, a CIR estimation unit estimates a group of impulse responses of a propagation path, a valid-impulse discriminator selects impulse responses (CIR), which are greater than a predetermined threshold value, from the impulse-response group, and a propagation path estimation unit generates a matrix expression using a CIR estimation vector that includes the selected CIRs as elements, a matrix S, which is decided based upon number N of points of an IFFT used in OFDM modulation and number Nc of subcarriers used in actual transmission, and a propagation-path response vector, obtains the propagation-path response vector by solving this matrix expression and estimating the propagation path.

BACKGROUND OF THE INVENTION

This invention relates to a method and apparatus for estimating apropagation path. More particularly, the invention relates to apropagation path estimation method and apparatus for estimating apropagation path traversed by a transmit signal in a receiver whencommunication utilizing OFDM (Orthogonal Frequency DivisionMultiplexing) is performed.

Frequency-selective fading ascribable to a multipath environment occursin wideband wireless communications. An effective method of dealing withthis is multicarrier modulation, which divides the transmissionbandwidth into narrow bands (subcarriers) that do not undergofrequency-selective fading, and transmits the subcarriers in parallel.

At present, specifications regarding digital TV and audio broadcasts (inJapan and Europe) and wireless LAN (IEEE 802.11a) are being standardizedbased upon OFDM transmission, which is one type of multicarriermodulation. An OFDM-based modulation scheme has been proposed fornext-generation mobile communication systems as well.

With a wireless communications system that employs OFDM-basedmodulation, it is necessary to estimate the propagation pathcharacteristics (propagation path information) of all subcarriers. Theprecision of the estimation has a major effect upon transmission errorrate in a manner similar to that of other wireless communicationssystems that use coherent detection. For this reason, a wirelesscommunications system using OFDM-based modulation transmits a knownsymbol on a subcarrier used in transmission and estimates propagationpath information subcarrier by subcarrier. As mentioned above, theprecision of propagation path estimation has a major effect upon thetransmission error rate and hence there are many cases where use is madeof a technique that suppresses background noise contained in apropagation path estimation value estimated using a known symbol, or aso called pilot symbol. For example, a first prior-art technique is toaverage frequency between adjacent subcarriers [see Hiroyuki Atarashi,Sadayuki Abeta and Mamoru Sawahashi, “Performance of Forward LinkBroadband Packet TD-OFCDM with Iterative Channel Estimation”, TechnicalReport of IEICE., DSP2000-154, SAT2000-110, RSC2000-186 (2001-01)], anda second prior-art technique is forced zero substitution of animpulse-response group on an estimated propagation path (seeJP2000-341242).

The first prior-art technique performs averaging between adjacentsubcarriers utilizing coherence (uniformity) in the frequency direction,thereby suppressing background noise. For example, if we let h₁ to h₅₁₂represent the propagation path characteristics of 512 subcarriers, asshown in FIG. 32, the propagation path characteristics of three adjacentsubcarriers are averaged and the average is adopted as the propagationpath characteristic of the middle subcarrier. The first prior-arttechnique utilizes a certain property, namely that if the propagationpath characteristics in a coherent bandwidth that is proportional to thereciprocal of delay spread are coherent and M-number of subcarriersexist in this coherent bandwidth, then the propagation pathcharacteristics of these M-number of subcarriers will be the same. Thefirst prior-art technique is such that if the delay spread is small, theamount of fluctuation in the propagation path characteristics along thefrequency direction is slight (correlation is large) and thereforebackground noise can be suppressed effectively by increasing the numberof averaging operations in the frequency direction. With regard to thedefinition of delay spread, a difference develops between the arrivaltimes of received waves in a multipath environment. The spread betweenthese delay times is referred to as delay spread.

In the first prior-art technique, however, correlation between amountsof channel fluctuation between adjacent subcarriers diminishes as delayspread increases. Consequently, a problem which arises is thatestimation precision declines if the number of averaging operationsalong the frequency direction is made greater than necessary. Actualdelay spread involves a great deal of fluctuation and, in an outdoorenvironment, can be 0.2 to 2.0 μs in urban areas and 10 to 20 μs inmountainous areas. This means that with the first prior-art technique,it is necessary to select the optimum number of averaging operationswhile measuring delay spread. Further, even if the optimum number ofaveraging operations has been selected, a problem which arises is thataveraging cannot be performed in an environment where the delay spreadis large, as in mountainous areas, and background noise will not besuppressed and receiver performance will be degraded compared to thewithout-averaging technique.

The second prior-art technique compares the power of an impulse-responsegroup on an estimated propagation path with a predetermined thresholdvalue and forcibly substitutes zero for impulses that are below thethreshold value, thereby suppressing background noise. An OFDM signal issuch that a signal that has been mapped to a subcarrier is transmittedupon being converted to the time domain by IFFT processing. However, ifthe IFFT size (N-point IFFT) and number (Nc) of subcarriers used insignal transmission differ, this is equivalent to performingmultiplication by a rectangular window on the frequency axis. As aresult, a time signal in OFDM is a signal of a convoluted time responsefunction decided by the number (Nc) of subcarriers used. If thesubcarriers at the edges of the spectrum is not used for transmission,time response is followed by a sinc function. Under the condition thattime response is the sinc function, the second prior-art techniqueutilizes this feature to set the threshold value to a value that isapproximately 13 dB below the main lobe, thereby arranging it so that aside lobe of the sinc function will not be discriminated as a valid path(impulse).

If N-point IFFT processing is executed with N items of data serving asthe components of N-number of subcarrier components f₁ to f_(N), thefrequency spectrum is as indicated at (A) in FIG. 33. In OFDM, a signalthat has undergone IFFT processing is converted to an analog signal,baseband signal components of f₁ to f_(N) are extracted from the analogsignal by a low-pass filter, and these are up-converted to radiofrequency and transmitted. In order to select baseband signal componentsof f₁ to f_(N), a low-pass filter having a sharp cut-off characteristicis necessary. Fabricating such a filter, however, is difficult.Accordingly, carriers on both sides of the N-number of subcarriers f₁ tofN are not used in data transmission, i.e., Nc-number (Nc<N) ofsubcarriers are used in data transmission, as illustrated at (B) in FIG.33. When the number Nc of subcarriers used in data transmission and theIFFT size (=N) thus differ, the propagation-path response becomes a sincfunction and not an impulse and the peak value of the main lobediminishes to Nc/N, as illustrated in FIG. 34. Consequently, in a casewhere Nc=N holds, the propagation-path response becomes an impulse, asillustrated at (A) in FIG. 35, but if Nc<N Holds, it becomes a waveformon which the sinc function has been superimposed, as indicated at (B) inFIG. 35. The second prior-art technique sets the threshold value to avalue that is approximately 13 dB below the main lobe, therebysuppressing background noise in such a manner that a side lobe of thesinc function will not be discriminated as a valid path (impulse).

In the second prior-art technique, the side lobes of the sinc functionare eliminated and only the main lobe is discriminated as a valid path.However, since the amplitude of the main lobe diminishes to Nc/N owingto the nature of the sinc function, a problem with the second prior-arttechnique is a residual estimation error. Further, in propagationenvironment in which path spacing is small, interference developsbetween the side lobes of the sinc function, the combined value in theoverlapped sample exceeds the threshold value and a path is erroneouslyjudged to be present where no path exists.

SUMMARY OF THE INVENTION

Accordingly, an object of the present invention is to provide apropagation path estimation method and apparatus in which it is possibleto suppress background noise irrespective of the propagationenvironment, such as delay spread and path spacing.

A further object of the present invention is to provide a propagationpath estimation method and propagation path estimation apparatus capableof correctly estimating propagation paths and improving BERcharacteristics, even when path positions in a real multipathenvironment deviate from measured sample positions.

The present invention provides a propagation path estimation method andapparatus of a receiver in an OFDM (Orthogonal Frequency DivisionMultiplexing) communication system for performing communication by OFDM.

A first propagation path estimation method according to the presentinvention comprises the steps of: estimating an impulse-response groupof a propagation path; selecting impulse responses, which are greaterthan a predetermined threshold value, from the impulse-response group;substituting zero for samples other than a prescribed number of samplesbracketing a maximum peak in the impulse responses selected; andestimating the propagation path using the impulse responses obtained bysubstitution.

A second propagation path estimation method according to the presentinvention comprises the steps of: estimating an impulse-response groupof a propagation path; selecting propagation-path impulse responses(CIRs), which are greater than a predetermined threshold value, from thepropagation-path impulse-response group; and generating a matrixexpression using a CIR estimation vector({circumflex over ({overscore (h)})}_(CIR))that includes the selected CIRs as elements, a matrix S, which isdecided by number N of points of IFFT used in OFDM modulation and numberNc of subcarriers used in actual transmission, and a propagation-pathresponse vector(ĥ_(t))and obtaining the propagation-path response vector by solving thismatrix expression. The matrix expression is{circumflex over ({overscore (h)})} _(CIR) =S·{overscore (h)} _(t) +P_(t) *·{overscore (w)}(where P_(t)* is a conjugate transposed matrix of known pilot symbols)taking a noise power vector {right arrow over (w)} into account. Thepropagation-path response vector is found from this matrix expression.

Further, the matrix S is a sinc function matrix decided by the number Nof points of the IFFT and number Nc of subcarriers, an inverse matrix ofthe S matrix is found, the inverse matrix is used to multiply the CIRestimation vector to thereby calculate the propagation-path responsevector, and those elements of the propagation-path response vectorobtained by calculation that are less than a threshold value are madezero to estimate the propagation path. Alternatively, the matrix S is asinc function matrix decided by the number N of points of the IFFT andnumber Nc of subcarriers, a weight matrix that is in accordance with theminimum mean square error (MMSE) is obtained using the matrix S andnoise variance, this matrix is used to multiply the CIR estimationvector to thereby calculate the propagation-path response vector, andthe propagation path is estimated from the propagation path vector.

A third propagation path estimation method of this invention has thesteps of estimating impulse responses in the frequency domain of apropagation path; M-fold oversampling (where M is an integer greaterthan or equal to 1) of estimated impulse responses; converting M-foldoversampled impulse responses into the time domain; selecting an impulseresponse equal to or greater than a predetermined threshold value, fromamong the time-domain impulse responses; replacing everything other thana prescribed number of samples before and after the maximum peak in theselected impulse response with a prescribed value; estimating the timeresponse of the propagation path, using the impulse response obtained bythe above replacement; and converting the estimated time response intothe frequency domain, performing M-fold down sampling, and thenestimating the M-fold downsampled propagation path.

A fourth propagation path estimation method of this invention has thesteps of estimating impulse responses in the frequency domain of apropagation path; M-fold oversampling (where M is an integer greaterthan or equal to 1) of estimated impulse responses; converting M-foldoversampled impulse responses into the time domain; selecting an impulseresponse equal to or greater than a predetermined threshold value, fromamong the time-domain impulse responses, and of generating an impulseresponse vector; creating a time response function matrix according totime response functions, based on the number N of IFFT points used inOFDM modulation and on the number N_(C) of sub-carriers used in actualpropagation, and of multiplying the inverse matrix of the above matrixby the above impulse response vector to estimate the propagation pathtime response; and converting the estimated time response into thefrequency domain, and then of performing M-fold downsampling andestimating the propagation path.

A first propagation path estimation apparatus according to the presentinvention comprises: a CIR estimation unit for estimating a group ofimpulse responses of a propagation path; a valid-impulse discriminatorfor selecting impulse responses, which are greater than a predeterminedthreshold value, from the impulse-response group and substituting zerofor samples other than a prescribed number of samples bracketing amaximum peak in the impulse responses selected; and a propagation pathestimation unit for estimating the propagation path using the validimpulse responses.

A second propagation path estimation apparatus according to the presentinvention comprises: a CIR estimation unit for estimating an impulseresponse (CIR) group of a propagation path; a valid-impulsediscriminator for selecting propagation-path impulse responses (CIR),which are greater than a predetermined threshold value, from thepropagation-path impulse-response group; and a propagation pathestimation unit for generating a matrix expression using a CIRestimation vector({circumflex over ({overscore (h)})}_(CIR))that includes the selected CIRs as elements, a matrix S, which isdecided by a number N of points of an IFFT used in OFDM modulation andnumber Nc of subcarriers used in actual transmission, and apropagation-path response vector({overscore (h)}_(t))and obtaining the propagation-path response vector and estimating thepropagation path by solving this matrix expression.

A third propagation path estimation apparatus of the present inventioncomprises an impulse response estimation unit, which estimates theimpulse response in the frequency domain of propagation paths; anoversampling unit, which performs M-fold (where M is an integer greaterthan or equal to 1) oversampling of estimated impulse responses; aninverse Fourier transform unit, which converts M-fold oversampledimpulse responses into the time domain; a valid impulse judgment unit,which selects an impulse response equal to or greater than apredetermined threshold value, from among the time-domain impulseresponses; an estimation unit, which replaces everything other than aprescribed number of samples before and after the maximum peak in theselected impulse response with a prescribed value and estimates the timeresponse of the propagation path; a Fourier transform unit, whichconverts the estimated propagation path time response into the frequencydomain; and, a propagation path estimation unit, which performs M-folddownsampling of the time response in the frequency domain and estimatesthe propagation path.

A fourth propagation path estimation apparatus of the present inventioncomprises an impulse response estimation unit, which estimates theimpulse response in the frequency domain of propagation paths; anoversampling unit, which performs M-fold (where M is an integer greaterthan or equal to 1) oversampling of estimated impulse responses; aninverse Fourier transform unit, which converts M-fold oversampledimpulse responses into the time domain; a valid impulse judgment unit,which selects an impulse response equal to or greater than apredetermined threshold value, from among the time-domain impulseresponses, and generates an impulse response vector; a propagation pathtime response estimation unit, which creates a time response functionmatrix using time response functions, based on an integral multiple M×Nof the number N of IFFT points used in OFDM modulation and on the numberN_(C) of sub-carriers used in actual propagation, and which estimatesthe propagation path time response by multiplying the inverse matrix ofthe above matrix by the above impulse response vector; a Fouriertransform unit, which converts the estimated propagation path timeresponse into the frequency domain; and means of performing M-folddownsampling of the frequency domain propagation path time response andfor estimating the propagation path.

In accordance with the present invention applied to communication usingOFDM-based modulation in which subcarriers not used in data transmissionexist, even if a delayed wave that exceeds a GI (Guard Interval) isgenerated, it is possible to obtain a propagation path estimation valuein which background noise is suppressed to a level equivalent to that ina case where the propagation path is known.

Further, in accordance with the present invention, it is possible tosuppress background noise irrespective of the propagation environmentsuch as delay spread and path spacing.

According to this invention, even when path positions in an actualmultipath environment deviate from sampled positions measured by thesystem, error due to deviation from the sampled positions can besuppressed through oversampling. As a result, background noise can alsobe suppressed, and by downsampling after noise suppression thepropagation path time response characteristics for paths can becorrectly estimated, and BER characteristics can be improved.

Other features and advantages of the present invention will be apparentfrom the following description taken in conjunction with theaccompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram illustrating the configuration of an OFDMcommunication system having a propagation path estimation unit accordingto the present invention;

FIG. 2 is a diagram useful in describing a data format and aserial-to-parallel conversion of an S/P converter;

FIG. 3 is a diagram useful in describing insertion of a guard interval;

FIG. 4 is a block diagram of a propagation path estimation unit;

FIG. 5 is a diagram useful in describing a propagation-path responsevector;

FIG. 6 is a diagram useful in describing a receive-signal vector in acase where N=8 holds;

FIG. 7 is a diagram useful in describing the waveform of a sincfunction;

FIG. 8 is a diagram useful in describing the column vector of an Smatrix;

FIG. 9 is a diagram useful in describing the CIR elements of a CIRestimation vector;

FIG. 10 is another block diagram of a propagation path estimation unit;

FIG. 11 is a diagram useful in describing the operation of adiscriminator for re-discriminating impulses;

FIG. 12 is a flowchart of propagation path estimation processingexecuted by the propagation path estimation unit;

FIG. 13 is a block diagram illustrating a weight generator forcalculating weight based upon the MMSE method;

FIG. 14 is a block diagram illustrating a second embodiment of an OFDMcommunication system having a propagation path estimation unit accordingto the present invention;

FIG. 15 is a block diagram illustrating a third embodiment in which apropagation path estimation unit according to the present invention isapplied to path search of a RAKE receiver;

FIG. 16 is a diagram useful in describing first simulation parameters;

FIG. 17 illustrates an Eb/N0 vs. MSE characteristic, which is the resultof a first simulation;

FIG. 18 is a diagram useful in describing second simulation parameters;

FIG. 19 illustrates a delay spread vs. required Eb/N0 characteristic,which is the result of a first simulation;

FIG. 20 is a block diagram of the OFDM communication system of a fourthembodiment;

FIG. 21 is an OFDM frame format example;

FIG. 22 explains the relation between the number N of IFFT and FFTpoints and the number Nc of subcarriers used in actual data propagation,and the relation between Nc data items and IFFT data input terminals;

FIG. 23 shows the block configuration of the propagation path estimationunit in the fourth embodiment;

FIG. 24 explains the output of the N-point Fourier transform unit;

FIG. 25 explains M-fold oversampling and downsampling;

FIG. 26 explains zero padding;

FIG. 27 explains the path model used in simulations;

FIG. 28 shows the Eb/No versus BER characteristics for channel A whenusing 16QAM;

FIG. 29 shows the Eb/No versus BER characteristics for channel B whenusing 16QAM;

FIG. 30 shows the Eb/No versus BER characteristics for channel A whenusing 64QAM;

FIG. 31 shows the Eb/No versus BER characteristics for channel B whenusing 64QAM;

FIG. 32 is a diagram useful in describing a first prior-art techniquefor suppressing background noise by performing averaging betweenadjacent subcarriers;

FIG. 33 illustrates a frequency spectrum in a case where all subcarriersf₁ to f_(N) are used in data transmission and a case where data istransmitted using subcarriers obtained by deleting subcarriers on bothsides in the prior art;

FIG. 34 is a diagram useful in describing propagation-path response (asinc function) in a case where subcarriers on both sides of N-number ofsubcarriers f₁ to f_(N) are not used in data transmission in the priorart; and

FIG. 35 is a diagram useful in describing problems encountered with asecond prior-art technique.

DESCRIPTION OF THE PREFERRED EMBODIMENTS (A) First Embodiment

In execution of propagation path estimation of an OFDM receiver in anOFDM communication system, a CIR (Channel Impulse Response) estimationunit estimates an impulse-response group of a propagation path, avalid-impulse discriminator selects impulse responses (CIR), which aregreater than a predetermined threshold value, from the impulse-responsegroup, and a propagation path estimation unit generates a matrixexpression using a CIR estimation vector that includes the selected CIRsas elements, a matrix S, which is decided by a number N of points of anIFFT used in OFDM modulation and number Nc of subcarriers used in actualtransmission, and a propagation-path response vector, and obtains thepropagation-path response vector and estimates the propagation path bysolving this matrix expression.

FIG. 1 is a block diagram illustrating the configuration of an OFDMcommunication system having a channel estimation unit according to thepresent invention. An OFDM transmitter 10 includes an encoder 11 forencoding binary data by, e.g., convolutional encoding or turbo encoding,and a modulator 12 for modulating the encoded data by, e.g., QPSK, afterinterleaving is performed. A serial/parallel (S/P) converter 13 convertsa modulated data symbol or pilot symbol to a parallel data sequence ofNc symbols and generates Nc-number of subcarrier components.

The OFDM transmitter 10 further includes an N-point inverse fast-Fouriertransform (IFFT) unit 14 that applies inverse fast-Fourier transform(IFFT) processing to the Nc-number of subcarrier components (modulateddata), which enters from the S/P converter 13, substituting zero for(N-Nc)-number of subcarriers of the N-number of subcarriers. Aparallel/serial (P/S) converter 15 converts, to serial data, N-number ofitems of time-series data obtained by the IFFT processing and outputsthe serial data as an OFDM symbol. The transmitter further includes aguard-interval insertion unit 16 that inserts a guard interval GI intothe OFDM symbol comprising the N-number of items of time-series data; adigital/analog (D/A) converter 17 that converts the signal, which isoutput from the guard-interval insertion unit 16, to an analog signal; alow-pass filter 18 for selecting and outputting a baseband signalcomponent; and a radio unit 19 for up-converting the baseband signal toa radio frequency, subsequently amplifying the signal and transmittingit from an antenna ATS. The signal that has been transmitted from theantenna ATS is transmitted over a multipath propagation path (multipathfading channel) 20 and is received by an OFDM receiver 30. AWGN(Additive White Gaussian Noise) is impressed upon the transmit signalduring propagation.

FIG. 2 is a diagram useful in describing a data format and theserial-to-parallel conversion performed by the S/P converter 13. Hereone frame is composed of 32×Nc symbols in which a pilot P has beentime-multiplexed to the forward end of transmit data DT. The pilot P perframe is composed of, e.g., 4×Nc symbols and the transmit data iscomposed of 28×Nc symbols. The S/P converter 13 outputs Nc symbols ofthe pilot the first four times as parallel data and subsequently outputsNc symbols of the transmit data 28 times as parallel data. As a result,an OFDM symbol comprising four pilot symbols can be transmitted in oneframe interval, the propagation path (channel) can be estimated on thereceiving side using these pilot symbols and channel compensation(fading compensation) becomes possible.

FIG. 3 is a diagram useful in describing insertion of a guard interval.If an IFFT output signal conforming to N-number of subcarrier samples(=1 OFDM symbol) is adopted as one unit, insertion of the guard intervalsignifies copying the tail-end portion of the signal to the leading endthereof. By inserting a guard interval GI, it is possible to eliminatethe effects of intersymbol interference (ISI) caused by multipath.

With reference again to FIG. 1, the OFDM receiver 30 includes a bandpassfilter (BPF) 31 that removes unwanted frequency components by applyingfiltering to the signal received by an antenna ATR; a downconverter(D/C) 32 for frequency-converting the radio signal to a basebandfrequency; an analog/digital converter (not shown) for converting theanalog baseband signal to digital data; a guard-interval removal unit 33for removing a guard interval; and an S/P converter 34 for convertingN-number of items of time-series data, from which the guard interval hasbeen removed, to parallel data and inputting a receive-signal vector({overscore (R)}_(t))to a propagation path estimation unit 35 and propagation pathcompensator 36. By way of a method (described later) using pilotsymbols, the propagation path estimation unit 35 calculates thefollowing propagation-path response vector comprising N-number of itemsof time-series elements:({overscore (h)}_(t))The propagation path compensator 36 multiplies the N-number of items oftime-series data of the receive-signal vector({overscore (R)}_(t))by each element of the following propagation-path response complexconjugate vector:({overscore (h)}_(t)*)

An N-point Fourier transform unit 37 applies N-point FFT processing toN-number of items of time-series data that has undergone propagationpath compensation, thereby outputting Nc-number of subcarriercomponents. A P/S converter 38 outputs the Nc-number of subcarriercomponents serially in order, a demodulator 39 demodulates the inputsignal by, e.g., QPSK, and a decoder 40 decodes the input data afterdeinterleaving is performed and outputs the decoded signal.

(a) First Configuration of Propagation Path Estimation Unit

FIG. 4 is a block diagram of the propagation path estimation unit 35.The propagation path estimation unit 35 includes a CIR estimator 51 forestimating propagation-path impulse response (channel impulse response)CIR using the receive-signal vector({overscore (R)}_(t))and a pilot-signal vector({overscore (P)}_(t))and outputting a CIR estimation vector({circumflex over ({overscore (h)})}_(CIR))The propagation path estimation unit further includes a valid-impulsediscriminator 52 that compares each CIR element of the CIR estimationvector ({circumflex over ({overscore (h)})}_(CIR)) with a thresholdvalue TH1, maintains CIR elements above the threshold value TH1 andmakes zero the CIR elements below the threshold value TH1 (i.e., makesthese CIR elements non-existent); a column vector generator 53 forgenerating column vectors of an S matrix (described later) using Nc andN; an S-matrix generator 54 for generating an S matrix using the columnvectors; a weight generator 55 for calculating an inverse matrix S⁻¹ ofthe S matrix as a weight matrix X; and a multiplier 56 for multiplyingthe CIR estimation value by a weight and outputting a propagation-pathresponse vector({overscore (h)}_(t))The operation of each of these units will be described usingmathematical expressions.

(a-1) Calculation of CIR Estimation Vector

In communication using an OFDM transmission scheme, pilot signals (pilotsymbols) that usually have equal power are disposed in the frequencydomain and CIR is estimated using these pilot signals. A signal vector({overscore (P)}_(f))in the frequency domain of a pilot signal and a signal vector({overscore (P)}_(t))in the time domain are written as follows, respectively:{overscore (P)} _(f) =[P _(f)(0)P _(f)(1) . . . P _(f)(N−1)]^(T)  (1){overscore (P)} _(t) =[P _(t)(0)P _(t)(1) . . . P _(t)(N−1)]^(T)  (2)where T represents a transposed matrix. The power of each element of{overscore (P)}_(f)takes on a value of 0 or 1. Specifically, N represents the IFFT size.The power of Nc-number of subcarrier signals that transmit a pilotsignal is 1, and the power of (N-Nc)-number of subcarrier signals thatdo not transmit a pilot signal is 0.

The relationship between{overscore (P)}_(f) and {overscore (P)}_(t)is as follows: $\begin{matrix}{{P_{t}(k)} = {{\mathcal{J}^{- 1}\left\{ {\overset{\_}{P}}_{f} \right\}} = {\frac{1}{N}{\sum\limits_{n = 0}^{N - 1}\quad{{P_{f}(n)}{\mathbb{e}}^{j\quad 2\pi\quad{{kn}/N}}}}}}} & (3)\end{matrix}$whereℑ⁻¹indicates IFFT processing of a number N of samples.

The true propagation-path response vector({overscore (h)}_(t))in the time domain and additive noise({overscore (w)})are written as follows, respectively:{overscore (h)} _(t) =[h _(t)(0)h _(t)(1) . . . h _(t)(N−1)]^(T)  (4){overscore (w)}=[w _(t)(0)w _(t)(1) . . . w _(t)(N−1)]^(T)  (5)If it is assumed that eight propagation paths PT₀ to PT₇ (multipath)exist between the transmitter 10 and receiver 30, as illustrated at (A)in FIG. 5, then({overscore (h)}_(t))will be the propagation-path response vector of each path. Thispropagation-path response vector is illustrated at (B) in FIG. 5, by wayof example. For the sake of simplicity, it is assumed that there is nointersymbol interference and therefore the guard interval (GI) of thepilot signal will be described as having a length greater than the datasymbol length.

The receive-signal vector({overscore (R)}_(t))in the time domain can be expressed as follows: $\begin{matrix}{{\overset{\_}{R}}_{t} = \left\lbrack {{R_{t}(0)}{R_{t}(1)}\quad\cdots\quad{R_{t}\left( {N - 1} \right)}} \right\rbrack^{T}} & (6) \\{{R_{t}(k)} = {{\sum\limits_{n = 0}^{N - 1}\quad{{h_{t}(n)}{P_{t}\left( {k - n} \right)}}} + {w(k)}}} & (7)\end{matrix}$

FIG. 6 illustrates a case where N=8 holds. Additive noise, however, isignored. The uppermost sequence in FIG. 6 is a time-series signal thatarrives at the receiver 30 via path PT₀, the second sequence is atime-series signal that arrives at the receiver 30 via path PT₁, and an(i+1)th sequence is a time-series signal that arrives at the receiver 30via path PT_(i). Further, Pt(j) represents a pilot, Dt(j) denotes data,1 signifies a signal in the time domain of the pilot signal, 2 the GI ofthe pilot signal, 3 a data signal that follows the pilot signal, and 4the GI of the data signal.

If the GI of the pilot signal is greater than the length of the datasymbol, as mentioned above, then cyclic convolution is assured in thereceive signal within the FFT window. For this reason, thereceive-signal vector({overscore (R)}_(t))can be expressed as the following matrix: $\begin{matrix}{\left\lfloor \begin{matrix}{R_{t}(0)} \\{R_{t}(1)} \\\cdots \\{R_{t}\left( {N - 1} \right)}\end{matrix} \right\rfloor = {{\left\lfloor \begin{matrix}{P_{t}(0)} & {P_{t}\left( {N - 1} \right)} & \cdots & {P_{t}(1)} \\{P_{t}(1)} & {P_{t}(0)} & \cdots & {P_{t}(2)} \\\cdots & \cdots & \cdots & \cdots \\{P_{t}\left( {N - 1} \right)} & {P_{t}\left( {N - 2} \right)} & \cdots & {P_{t}(0)}\end{matrix} \right\rfloor \cdot \left\lfloor \begin{matrix}{h_{t}(0)} \\{h_{t}(1)} \\\cdots \\{h_{t}\left( {N - 1} \right)}\end{matrix} \right\rfloor} + \left\lfloor \begin{matrix}{w_{t}(0)} \\{w_{t}(1)} \\\cdots \\{w_{t}\left( {N - 1} \right)}\end{matrix} \right\rfloor}} & (8)\end{matrix}$This receive-signal vector is input to the CIR estimator 51 of FIG. 4.

The channel impulse response CIR can be estimated, as indicated below,from Equation (8) by sliding correlation using the receive-signal vector({overscore (R)}_(t))and known pilot-signal vector({overscore (P)}_(t))Specifically, the CIR estimation vector becomes as follows:{circumflex over ({overscore (h)})} _(CIR) =[ĥ _(CIR)(0)ĥ _(CIR)(1) . .. ĥ _(CIR)(N−1)]^(T)  (9) $\begin{matrix}{{{\hat{h}}_{CIR}(k)} = {\sum\limits_{n = 0}^{N - 1}\quad{{R_{t}(n)}{P_{t}^{*}\left( {k - n} \right)}}}} & (10)\end{matrix}$The upper equation becomes as follows when expressed by a matrix:$\begin{matrix}{\left\lfloor \begin{matrix}{{\hat{h}}_{CIR}(0)} \\{{\hat{h}}_{CIR}(1)} \\\cdots \\{{\hat{h}}_{CIR}\left( {N - 1} \right)}\end{matrix} \right\rfloor = {\left\lfloor \begin{matrix}{P_{t}^{*}(0)} & {P_{t}^{*}(1)} & \cdots & {P_{t}^{*}\left( {N - 1} \right)} \\{P_{t}^{*}\left( {N - 1} \right)} & {P_{t}^{*}(0)} & \cdots & {P_{t}^{*}\left( {N - 2} \right)} \\\cdots & \cdots & \cdots & \cdots \\{P_{t}^{*}(1)} & {P_{t}^{*}(2)} & \cdots & {P_{t}^{*}(0)}\end{matrix} \right\rfloor \cdot \left\lfloor \begin{matrix}{R_{t}(0)} \\{R_{t}(1)} \\\cdots \\{R_{t}\left( {N - 1} \right)}\end{matrix} \right\rfloor}} & (11)\end{matrix}$This can be transformed as follows:{circumflex over ({overscore (h)})} _(CIR) =P _(t) *·{overscore (R)}_(t) =P _(t) ^(*)·(P _(t) ·{overscore (h)} _(t) +{overscore (w)})=P _(t)*·P _(t) ·{overscore (h)} _(t) +P _(t) *·{overscore (w)}  (12)where P_(t) is written as follows: $\begin{matrix}{P_{t} = \left\lfloor \begin{matrix}{P_{t}(0)} & {P_{t}\left( {N - 1} \right)} & \cdots & {P_{t}(1)} \\{P_{t}(1)} & {P_{t}(0)} & \cdots & {P_{t}(2)} \\\cdots & \cdots & \cdots & \cdots \\{P_{t}\left( {N - 1} \right)} & {P_{t}\left( {N - 2} \right)} & \cdots & {P_{t}(0)}\end{matrix} \right\rfloor} & (13)\end{matrix}$

The CIR estimator 51 estimates the CIR estimation vector({circumflex over ({overscore (h)})}_(CIR))from Equation (11) and inputs this to the valid-impulse discriminator52.

The foregoing has been described with regard to a CIR estimation methodin the time domain, though similar processing is possible also in thefrequency domain. That is, in the frequency domain, frequency-domainsignal processing is executed using the following equation:${{\hat{h}}_{CIR}(k)} = {{F^{- 1}\left\{ \frac{R_{f}(k)}{P_{f}(k)} \right\}} = {F^{- 1}\left\{ \frac{{R_{f}(k)}{P_{f}^{\phi}(k)}}{{{P_{f}^{\phi}(k)}}^{2}} \right\}}}$instead of Equation (10) of the time domain. Though 0≦k≦N−1 is the sameas in the time domain, here k is a subcarrier number (k is a samplenumber in the time domain). In order to eventually execute IFFT in theprocessing of the frequency-domain signal of the above equation, thefrequency signal enclosed by { } is converted to a time-domain signaland becomes the time-domain signal of Equation (10). Executing signalprocessing in the frequency domain is more advantageous than directcalculation in the time domain in that the standard of the circuitry issmaller.

(a-2) Discrimination of Valid Impulses

Each element S_(ij) of P_(t)*·P_(t) in the CIR estimation vector({circumflex over ({overscore (h)})}_(CIR)) can be expressed as follows:$\begin{matrix}{S_{ij} = {\sum\limits_{n = 0}^{N - 1}\quad{{P_{t}^{*}\left( {n - i} \right)} \cdot {P_{t}\left( {n - j} \right)}}}} & (14)\end{matrix}$

Furthermore, if a column is expressed as a vector (e.g., j=0), thenEquation (14) can be transformed to an equation of a cyclicconvolutional calculation as follows: $\begin{matrix}{{\overset{\_}{S}}_{0} = {\sum\limits_{n = 0}^{N - 1}\quad{{P_{t}^{*}\left( {n - i} \right)} \cdot {P_{t}(n)}}}} & (15)\end{matrix}$

A cyclic convolutional calculation is equivalent to what is obtained byapplying an IFFT to a product in a frequency device and therefore theabove can be transformed to{overscore (S)} ₀ℑ⁻¹ {ℑ{{overscore (P)} _(t) *}·ℑ{{overscore (P)} _(t)^(T)}}=ℑ⁻¹ {{overscore (P)} _(f) *·{overscore (P)} _(f) ^(T)}  (16)Hereℑ, ℑ⁻¹indicate FFT processing and IFFT processing, respectively. Because{overscore (P)}_(f) *·{overscore (P)} _(f) ^(T)is power of the pilot signal, this equation compensates for theinformation of each element of{overscore (P)}_(f)and signifies that an untransmitted subcarrier becomes 0 and that asubcarrier used in transmission becomes 1.

By thus compensating for the information of each element of{overscore (P)}_(f)the result will be that if an untransmitted subcarrier does not exist(i.e., if all elements of{overscore (P)}_(f) *·{overscore (P)} _(f) ^(T)are 1), then{overscore (S)}₀will be an impulse, and if an untransmitted subcarrier does exist (i.e.,if an element of{overscore (P)}_(f)*·{overscore (p)}_(f) ^(T)contains 0), then{overscore (S)}₀will become a sinc function specified by N, Nc. In other words, since anend subcarrier is made an untransmitted subcarrier (i.e., since thissubcarrier is not used in data transmission), the result is arectangular function on the frequency axis. If this is subjected to anIFFT, it becomes a sinc function on the time axis. Accordingly,{overscore (S)}₀is an even function and therefore can be expressed as follows:{overscore (S)} ₀ =[s(0)s(1) . . . s((N/2)−1)s(N/2)s((N/2)−1 . . .S(1)]^(T)  (17)

FIG. 7 is a diagram useful in describing the waveform of a sincfunction. A main lobe has a peak value A that is Nc/N and a width W thatbroadens as Nc decreases. In FIG. 7, items of time-series data S(0),S(1), . . . , S(1) that prevail when the left half of the sinc functionis folded over as indicated by the dashed line become the elements ofthe column vector{overscore (S)}₀in Equation (17). A kth vector is a vector that results when{overscore (S)}₀is shifted by k. Finally,S=P _(t) *·P _(t)can be expressed as the matrix indicated below. $\begin{matrix}{S = \left\lfloor \begin{matrix}{s(0)} & {s(1)} & \cdots & \cdots & \cdots & \cdots & {s(1)} \\{s(1)} & {s(0)} & \cdots & \cdots & \cdots & {s\left( {\left( {N/2} \right) -} \right.} & \cdots \\\quad & \quad & \quad & \quad & \quad & \left. 1 \right) & \quad \\\cdots & {s(1)} & \cdots & \cdots & \cdots & {s\left( {N/2} \right)} & {s\left( {\left( {N/2} \right) -} \right.} \\\quad & \quad & \quad & \quad & \quad & \quad & \left. 1 \right) \\{s\left( {\left( {N/2} \right) -} \right.} & \cdots & \cdots & \cdots & \cdots & \cdots & {s\left( {N/2} \right)} \\\left. 1 \right) & \quad & \quad & \quad & \quad & \quad & \quad \\{s\left( {N/2} \right)} & {s\left( {\left( {N/2} \right) -} \right.} & \cdots & \cdots & \cdots & {s(1)} & \cdots \\\quad & \left. 1 \right) & \quad & \quad & \quad & \quad & \quad \\\cdots & {s\left( {N/2} \right)} & \cdots & \cdots & \cdots & {s(0)} & {s(1)} \\{s(1)} & \cdots & \cdots & \cdots & \cdots & {s(1)} & {s(0)}\end{matrix} \right\rfloor} & (18)\end{matrix}$

Specifically, the column vectors{overscore (S)}₀,{overscore (S)}₁,{overscore (S)}₂, . . .in FIG. 18 are the result of shifting the sinc function of FIG. 7successively by a time difference Åt of each element of thepropagation-path response vector({overscore (h)}_(t))in the manner shown in FIG. 8.

In view of the description thus far, the CIR estimation vector({circumflex over ({overscore (h)})}_(CIR))can be expressed as follows:{circumflex over ({overscore (h)})} _(CIR) =P _(t)*·(P _(t) ·{overscore(h)} _(t) +{overscore (w)})=P _(t) *·P _(t) ·{overscore (h)} _(t) +P_(t) *·{overscore (w)}=S·{overscore (h)} _(t) +P _(t) *·{overscore(w)}  (19)

In other words, {circumflex over ({overscore (h)})}_(CIR) can beconsidered to be the result of adding additive noise to the productobtained by multiplying S by the propagation-path response vector{overscore (h)}_(t)HereSis formed by a vector having the shape of a sinc function and thereforetakes on a value that becomes sharply smaller as distance increases froms(0), which is the peak value of the main lobe. FIG. 9 is a diagramuseful in describing the CIR estimation vector in a case where{hacek over ({overscore (h)})} _(CIR) =S·{overscore (h)} _(t)holds, and it will be understood this takes on a value that becomessharply smaller as distance increases from s(0), which is the peak valueof the main lobe. Accordingly, the information (energy) of thepropagation path vector{overscore (h)}_(t)is dispersed from the main lobe to a certain specific interval.

In view of the above, the CIR estimation vector({circumflex over ({overscore (h)})}_(CIR))calculated according to Equation (11) has the shape shown in FIG. 9.Accordingly, the valid-impulse discriminator 52 compares each CIRelement (impulse) of the CIR estimation vector with the threshold valueTH1, selects impulses that are greater than the threshold value TH1 andmakes impulses below the threshold value non-existent (i.e. makes themzero). For example, assume that CIR element h_(CIR)(3)=0 holds in FIG. 9so that propagation-path response is rendered non-existent. Further, 0is substituted for samples other than a prescribed number m of samplesbracketing the maximum peak value s(0) of the selected impulse. Thethreshold value TH1 is made power that is lower than the maximum peakvalue of the CIR by a value decided based upon the number N of points ofthe IFFT used in OFDM modulation and the number Nc of subcarriers usedin actual transmission. Alternatively, the threshold value TH1 is madepower that is greater than background noise power, which has beenestimated by some method, by a predetermined value.

In other words, a set vector{overscore (l)}that indicates the sample position of an impulse response greater than athreshold value 1 is output to the weight generator 55 and{circumflex over ({overscore (h)})}_(CIR)is output as{circumflex over ({overscore (h)})}_(CIR)that is the result of degeneracy of size conforming to the set vector.The weight generator 55 generates and outputs the following weightmatrix from the set vector{overscore (l)}and m: $\begin{matrix}{X = \begin{bmatrix}0 & \quad & \quad & \quad & \quad & \quad & \quad & \quad \\\quad & 0 & \quad & \quad & \quad & \quad & 0 & \quad \\\quad & \quad & 1 & \quad & \quad & \quad & \quad & \quad \\\quad & \quad & \quad & \quad & ⋰ & \quad & \quad & \quad \\\quad & \quad & \quad & \quad & \quad & 1 & \quad & \quad \\\quad & 0 & \quad & \quad & \quad & \quad & 0 & \quad \\\quad & \quad & \quad & \quad & \quad & \quad & \quad & 0\end{bmatrix}} & \left( {19a} \right)\end{matrix}$

Here the interval of 1 is an index represented by the set vector({overscore (l)})and is m in number before and after. The multiplier 56 multiplies{circumflex over ({overscore (h)})}′_(CIR)and the weight matrix X together, whereby zero is substituted forsamples other than m-number of samples bracketing the selected impulseresponse.

As set forth above, the present invention is so adapted that each CIRelement (impulse) of the CIR estimation vector({circumflex over ({overscore (h)})}_(CIR))can be estimated independently, impulse responses greater than thethreshold value TH1 are selected and these impulses can be expressed bythe prescribed m-number of samples. As a result, the decline in energyis held to the minimum. This solves the problem encountered with theprior art, namely the inclusion of a permanent estimation error owing tothe fact that the amplitude of the main lobe is Nc/N. It should be notedthat the number m of samples is an already known value decided from thenumber N of points of the IFFT and the number Nc of subcarriers used inactual transmission.

(a-3) Generation of S Matrix, Generation of Weight and Calculation ofPropagation-Path Response Vector

In accordance with Equation (19), the propagation-path response vector{overscore (h)}_(t)can be obtained by multiplying the CIR estimation vector{circumflex over ({overscore (h)})}_(CIR)by the inverse matrix of S. That is, the propagation-path responsevector can be found according to the following equation:{circumflex over ({overscore (h)})} _(t) =S ⁻¹ ·{circumflex over({overscore (h)})} _(CIR) ={overscore (h)} _(t) +S ⁻¹ ·P _(t)*·{overscore (w)}  (20)

Accordingly, the column vector generator 53 generates the column vectors{overscore (S)}₀,{overscore (S)}₁,{overscore (S)}₂, . . .of the S matrix of Equation (18) using Nc and N, the S-matrix generator54 generates the S matrix using these column vectors, and the weightgenerator 55 calculates the inverse matrix S⁻¹ of the S matrix as theweight matrix. In accordance with Equation (20), the multiplier 56multiplies the CIR estimation vector{circumflex over ({overscore (h)})} _(CIR)(={circumflex over ({overscore(h)})} _(CIR))which is output from the valid-impulse discriminator 52, by the weightand outputs the propagation-path response vector({overscore (h)}_(t))It should be noted that since the sinc function is uniquely decided byN, Nc, the column vector generator 53 is capable of generating thecolumn vectors{overscore (S)}₀,{overscore (S)}₁,{overscore (S)}₂, . . .based upon the sinc function. In this case, the column vector generator53 previously calculates the column vectors{overscore (S)}₀,{overscore (S)}₁,{overscore (S)}₂, . . .that conform to Nc, N and stores the column vectors, whereby it can beso arranged that these column vectors are not calculated one by one.Further, according toS=P _(t) *·P _(t)the column vector generator 53 can calculate the column vectors{overscore (S)}₀,{overscore (S)}₁,{overscore (S)}₂, . . .by a matrix operation between the matrixP_(t)of Equation (13) and the conjugate transposed matrixP_(t)*and can input the column vectors to the S-matrix generator 54.

The valid-impulse discriminator 52 selects impulse responses above thethreshold value (TH1) from the CIR estimation vector{circumflex over ({overscore (h)})}_(CIR)and makes zero the impulse responses below the threshold value to makethem non-existent. The S-matrix generator 54 creates the S matrix makingzero the matrix elements that conform to the impulse responses that havebeen made zero, thereby reducing the size of the matrix and making itpossible to reduce calculations.

Thus, in accordance with the first propagation path estimation unit,each of the CIR elements of the CIR estimation vector{circumflex over ({overscore (h)})}_(CIR)can be estimated independently on a per-path basis. As a result, even ifpaths are close together and side lobes of the channel impulse responseCIR of each of the paths overlap each other, a CIR below the thresholdvalue TH1 can be discriminated correctly to be non-existent, therebymaking it possible to obtain a propagation path estimation value inwhich background noise has been suppressed.

Further, since CIR is estimated using a prescribed number of samplesfrom among the samples that constitute CIR above the threshold valueTH1, the CIR estimation error, which is a problem with the secondprior-art technique, can be reduced. Moreover, by making zero theunwanted side lobe portions, it is possible to perform propagation pathestimation in which background noise has been suppressed.

(b) Second Configuration of Propagation Path Estimation Unit

FIG. 10 is another block diagram of the propagation path estimation unit35. This arrangement differs from that of FIG. 4 in that a discriminator57 for re-discriminating impulses is provided. The discriminator 57compares the propagation-path response vector{overscore (h)}_(t)which has been obtained by calculating the inverse matrix, with athreshold value again, namely a threshold value TH2, makes zero theelements of the propagation-path response vector{overscore (h)}_(t)that are below the threshold value TH2 and outputs the results. Thereason for this is as follows: There are cases where the CIR estimationvector{circumflex over ({overscore (h)})}′_(CIR)selected by comparison with the threshold value in the valid-impulsediscriminator 52 contains an impulse that exceeds this threshold value(TH1) owing to the effects of the side lobes of the sinc function, asindicated by the dash-line arrow at (A) in FIG. 11. This impulse isoutput as a valid impulse from the valid-impulse discriminator 52, asindicated at (B) in FIG. 11. This impulse does not actually exist.Accordingly, the discriminator 57 compares the propagation-path responsevector{overscore (h)}_(t)from which the effects of the side lobes have been eliminated, asindicated (C) in FIG. 11, with a threshold value again, this time thethreshold value TH2, and substitutes zero for impulses that are belowthe threshold value TH2, as indicated at (D) in FIG. 11. As a result, inaccordance with the propagation path estimation unit of FIG. 10, it ispossible to suppress background noise further. It should be noted thatthe threshold value TH2 is decided in accordance with a criterionidentical with that of the threshold value TH1 described above.

FIG. 12 is a flowchart of propagation path estimation processingexecuted by the propagation path estimation unit 35. The CIR estimator51 estimates channel impulse response CIR using the receive-signalvector({overscore (R)}_(t))and pilot-signal vector({overscore (P)}_(t))and outputs the CIR estimation vector (step 101)({circumflex over ({overscore (h)})}_(CIR))The valid-impulse discriminator 52 compares each element of the CIRestimation vector({circumflex over ({overscore (h)})}_(CIR))with the threshold value TH1, maintains CIR elements of the CIRestimation vector that are greater than the threshold value TH1, andmakes zero the CIR elements below the threshold value TH1 (step 102).Next, the S-matrix generator 54 generates the S matrix (step 103). Theweight generator 55 calculates, as the weight matrix X, the matrixdecided by the set vector{overscore (l)}and m, or the inverse matrix S-1 of the S matrix (step 104). Themultiplier 56 multiplies the CIR estimation value by the weight andestimates the propagation-path response vector (step 105)({overscore (h)}_(t))The discriminator 57 compares the propagation-path response vector{overscore (h)}_(t)which has been obtained by calculating the inverse matrix, with athreshold value again, namely the threshold value TH2, makes zero theelements of the propagation-path response vector{overscore (h)}_(t)that are below the threshold value TH2 and outputs the results (step106).

(c) Reduction in Calculations

In order to reduce the amount of calculation, Equation (19) isdegenerated (S→S′) only to an index related to the set vector{overscore (l)}and it is possible to solve the inverse matrix. For example, in a casewhere impulses that have been selected by the valid-impulsediscriminator 52 can be put into block form, the inverse matrix is splitand generated block by block. This will be described for a case whereN=4 holds. With regard to the CIR estimation vector({circumflex over ({overscore (h)})}_(CIR))the following equation holds in view of Equation (19): $\begin{matrix}{\left\lfloor \begin{matrix}{{\hat{h}}_{CIR}(0)} \\{{\hat{h}}_{CIR}(1)} \\{{\hat{h}}_{CIR}(2)} \\{{\hat{h}}_{CIR}(3)}\end{matrix} \right\rfloor = {{\begin{bmatrix}{s(0)} & {s(1)} & {s(2)} & {s(1)} \\{s(1)} & {s(0)} & {s(1)} & {s(2)} \\{s(2)} & {s(1)} & {s(0)} & {s(1)} \\{s(1)} & {s(2)} & {s(1)} & {s(0)}\end{bmatrix} \cdot \begin{bmatrix}{h_{t}(0)} \\{h_{t}(1)} \\{h_{t}(2)} \\{h_{t}(3)}\end{bmatrix}} + {P_{t}^{*}\begin{bmatrix}{w(0)} \\{w(1)} \\{w(2)} \\{w(3)}\end{bmatrix}}}} & (21)\end{matrix}$

If we assume here that the CIR elements (impulses) of the CIR estimationvector that are greater than the threshold value are h_(CIR)(0) andh_(CIR)(1), then the above can be expressed by a matrix obtained bydegenerating the matrixSas follows: $\begin{matrix}{\left\lfloor \begin{matrix}{{\hat{h}}_{CIR}(0)} \\{{\hat{h}}_{CIR}(1)}\end{matrix} \right\rfloor = {{\begin{bmatrix}{s(0)} & {s(1)} \\{s(1)} & {s(0)}\end{bmatrix} \cdot \begin{bmatrix}{h_{t}(0)} \\{h_{t}(1)}\end{bmatrix}} + {P_{t}^{*}\begin{bmatrix}{w(0)} \\{w(1)}\end{bmatrix}}}} & (22)\end{matrix}$where the following holds: ${S^{\prime} = \begin{bmatrix}{S(0)} & {S(1)} \\{S(1)} & {S(0)}\end{bmatrix}},{{\overset{\_}{\hat{h}}}_{CIR}^{\prime} = \left\lfloor \begin{matrix}{{\hat{h}}_{CIR}(0)} \\{{\hat{h}}_{CIR}(1)}\end{matrix} \right\rfloor},{\overset{\_}{w} = \begin{bmatrix}{w(0)} \\{w(1)}\end{bmatrix}}$If Equation (21) can be transformed to Equation (22), then the amount ofcalculation for the inverse matrix can be reduced by a wide margin.Furthermore, even though the degenerate matrices become plural innumber, there is no correlation among them and therefore it will sufficeto solve the matrices individually.

The degenerate matrices thus obtained can be written as follows:{circumflex over ({overscore (h)})}′ _(CIR) =S′·{circumflex over({overscore (h)})} _(t) +P _(t) ′*{overscore (w)}′and therefore by multiplying by the inverse matrixS′the propagation-path response vector{overscore (h)}_(t)can be obtained as follows:{circumflex over ({overscore (h)})} _(t) =S′ ⁻¹ ·{circumflex over({overscore (h)})}′ _(CIR) ={overscore (h)} _(t) +S′ ⁻¹ ·P′ _(t)*{overscore (w)}′In other words, the matrix S decided by N and Nc is generated by theS-matrix generator 54, and the weight generator 55 generates S′ usingthe output {overscore (l)} of the valid-impulse discriminator 52 and Sfrom the S-matrix generator 54. Furthermore, the inverse matrix of S′ isfound and output to the multiplier 56 as the weight matrix X. Themultiplier 56 obtains the propagation-path response vector{overscore (h)}_(t)by multiplying{circumflex over ({overscore (h)})}′_(CIR)which is composed of the I component of{circumflex over ({overscore (h)})}_(CIR)and the weight matrix X together. Further, in a case where CIRs abovethe threshold value are only{overscore (h)}_(CIR)(0)or in a case where selected impulses are spaced apart, Equation (21)becomes as follows:ĥCIR(0)=s(0)·h _(t)(0)+P _(t)′(0)·w(0)  (23)In such case, it will suffice to divideĥ_(CIR)(0)by s(0) without finding the inverse matrix, thereby making it possibleto reduce the amount of calculation. Further, since s(0) is Nc/N, wehave the following: ${h_{t}(0)} = {{{\hat{h}}_{CIR}(0)}\frac{N}{Nc}}$This is equivalent to a solution to the problem of the prior art, namelythat the amplitude of the main lobe becomes Nc/N and results ininclusion of a permanent estimation error.

In summation, if propagation path impulse responses (CIRs) above athreshold value can be divided into a plurality of blocks, then thematrix expression is divided up block by block to obtain thepropagation-path response vector. In this case, if there is even asingle propagation path-impulse response (CIR) that belongs to theblocks, then the propagation-path response vector is found bymultiplying the CIR estimation vector by a fixed value, which is decidedby the number N of points of the IFFT and number Nc of subcarriers usedin signal transmission, without generating a matrix expression.

The amount of calculation involving the inverse matrix changes by a widemargin in dependence upon the number of CIR elements above the thresholdvalue in the CIR estimation vector({circumflex over ({overscore (h)})}_(CIR))Accordingly, the propagation path is estimated in accordance with amethod in which the amount of calculation is reduced in dependence uponthe state of the CIRs.

(d) Other Implementation of Weight Calculation

As set forth above, weight calculation is performed based upon theinverse of the S matrix and multiplication by the CIR estimation vector({circumflex over ({overscore (h)})}′_(CIR))is performed to calculate the propagation-path response vector{overscore (h)}_(t)However, in order to diminish the influence of additive noise, a methodof obtaining weight based upon the standard of MMSE (Minimum Mean SquareError) is conceivable. A method of calculating weight based upon theMMSE method will be described below.

If we write{circumflex over ({overscore (h)})} _(t) =X−{circumflex over ({overscore(h)})}′ _(CIR)then the weight matrixXthat minimizes the evaluation function $\begin{matrix}{J = {E\left\lfloor {{{\overset{\_}{\hat{h}}}_{t} - {\overset{\_}{h}}_{t}}}^{2} \right\rfloor}} & (24)\end{matrix}$is found by solving $\begin{matrix}{\frac{\partial J}{\partial X} = 0} & \left( {24a} \right)\end{matrix}$In Equation (24), J represents an evaluation function, E an expectedvalue,{circumflex over ({overscore (h)})}_(t)an estimated value and{overscore (h)}_(t)the actual propagation-path response vector. If Equation (24a) issolved, then the weight vector matrix is given by $\begin{matrix}{X = \frac{S^{*}}{{S}^{2} + {{{\overset{\_}{P}}_{t}^{*}}^{2} \cdot \sigma^{2} \cdot I}}} & (25)\end{matrix}$where∥·∥²represents the norm of the square of the vector, σ² the variance ofadditive noise and I a unit vector.

FIG. 13 is a block diagram illustrating a weight generator forcalculating weight based upon the MMSE method. A variance calculationunit 55 a calculates variance of the additive noise, and a weightcalculation unit 55 b calculates the weight matrix X according toEquation (25) and outputs the same. If the statistical properties ofnoise are understood, weight that takes noise into consideration can becalculated according to Equation (25) and a propagation-path responsevector{overscore (h)}_(t)of higher precision can be output.

(B) Second Embodiment

FIG. 14 is a block diagram illustrating a second embodiment of an OFDMcommunication system having a propagation path estimation unit accordingto the present invention. Components in FIG. 14 identical with those ofthe first embodiment of FIG. 1 are designated by like referencecharacters. This embodiment differ in the following points:

-   -   (1) A Fourier transform unit 41 is provided for applying FFT        processing to the time-domain propagation-path response vector        {circumflex over ({overscore (h)})}_(t)        that is output from the propagation path estimation unit 35, and        for outputting a propagation-path response vector in the        frequency domain        {circumflex over ({overscore (h)})}_(f)    -   (2) A propagation path compensator 42 is provided for performing        channel compensation by multiplying the Fourier-transformed        receive-signal vector        {overscore (R)} _(f)=[0 . . . 0R _(f)(0)R _(f)(1) . . . R        _(f)(N-1)0 . . . 0]^(T)        in the frequency domain by a propagation-path response complex        conjugate vector        {circumflex over ({overscore (h)})}*_(f)        i.e., based upon        R _(f)(i)·{circumflex over ({overscore (h)})}* _(f)0≦i≦N _(c)-1

(c) Third Embodiment

The foregoing relates to a case where the present invention is appliedto estimation of propagation path in an OFDM receiver. However, thepresent invention is also applicable to path search in a RAKE receiverof a CDMA (Code Division Multiplex Access) scheme. FIG. 15 is a blockdiagram illustrating a third embodiment in which the present inventionis applied to path search of a RAKE receiver.

A radio receiver 62 frequency-converts a high-frequency signal, whichhas been received by an antenna 61, to a baseband signal and appliesquadrature demodulation. A low-pass filter (LPF) 63 removes unwantedfrequency components and inputs the resultant signal to an AD converter64. The latter converts the input quadrature-demodulated signal todigital data, and a path searcher 65 searches a receive signal Rt forpaths constituting a multipath system. Fingers 66 a to 66 n are providedfor corresponding ones of paths of the multipath system and each has adespreader and a delay circuit. A timing generator 67 inputs the delaytimings of the paths to the despreaders of respective ones of thefingers 66 a to 66 n as despreading timings and inputs delay times,which are for matching the timings of the despread signals that areoutput from the despreaders, to the delay circuits of respective ones ofthe fingers 66 a to 66 n. A RAKE combiner 68 applies maximal ratiocombining to the despread signals that are output from the fingers 66 ato 66 n and outputs the combined signal to a channel codec (not shown),which is the next stage.

The path searcher 65, which has a structure identical with that of thepropagation path estimation unit of FIG. 4, calculates thepropagation-path response vector{overscore (h)}_(t)through the same method and performs a path search. This arrangementdiffers in that when a column vector of the S matrix is generated,impulse response of the low-pass filter 63 is used instead of{overscore (S)}₀This can be applied using impulse response values of a cosine roll-offfilter stipulated by standardization (3GPP) of CDMA, by way of example.That is, since a sinc function is uniquely decided by W_(LPF), W_(CDMA),the S matrix is created based upon the sinc function.

It should be noted that in all of the embodiments above, impulseresponses above a predetermined threshold value are selected from animpulse-response group of an estimated propagation path and zero issubstituted for samples other than a prescribed number of samples beforeand after the largest peak in the selected impulse responses. However,in an operation involving a floating decimal point or fixed decimalpoint, for example, it will suffice to use values in which the LSB ismade “1” and the other bits are made “0”, or values substantiallyhandled as zero as in a null state, etc.

Similarly, among elements of a propagation-path response vector obtainedby multiplying the CIR estimation vector by an inverse matrix of the Smatrix, elements below the threshold value are made zero in estimatingthe propagation path. However, in an operation involving a floatingdecimal point or fixed decimal point, for example, it will suffice touse values in which the LSB is made “1” and the other bits are made “0”,or values substantially handled as zero as in a null state, etc.

(D) Results of Simulation

The effects of the present invention will be made clear by simulationswhile comparing it with the second prior-art technique.

(a) First Simulation

A first simulation deals with a case where the number of paths of amultipath environment is small and path spacing is large. FIG. 16illustrates simulation parameters in the first simulation. In thetransmission model, the number of paths is two, the reception levels ofthe two paths are identical, the delay time between the two paths is 1sample per 300 samples, and the fading frequency is 960 Hz. Further, thethreshold value TH1 is −15 dB from the peak power in the CIR. This is avalue decided by Nc (=896) and N (=1024).

FIG. 17 illustrates an Eb/N0 vs. MSE (Mean Square Error) characteristicwithout background-noise suppression processing, according to the priorart and according to the present invention. The present inventionrelates to a case where the propagation-path response vector({overscore (h)}_(t))has been estimated according to Equation (22). In FIG. 17, the curve Ais the characteristic in a case where processing for suppressingbackground noise is not executed, the curve B is the characteristic in acase where the prior art is applied, and the curve C is thecharacteristic in a case where the present invention is applied.According to these characteristics, the prior art exhibits abackground-noise suppression capability in an environment where Eb/N0 ispoor. In an environment where Eb/N0 is favorable, however, a permanentestimation error is included and therefore no improvement incharacteristic is seen. With the present invention, on the other hand,the characteristic improves as Eb/N0 increases and therefore theinvention is effective in suppressing background noise. Further,simulation conditions indicate effectiveness even in a case where thepath spacing is very small or in an environment in which normalizeddelay spread is large (about 0.14).

(b) Second Simulation

A second simulation deals with a case where the number of paths of amultipath environment is large and path spacing is small. This indicatesthat the present invention is effective even in a transmissionenvironment in which paths are close together. FIG. 18 illustratessimulation parameters in the second simulation. In the transmissionmodel, the number of paths is 12, the reception levels of the pathsdiminish exponentially in accordance with delay time, the path spacingis one sample and the fading frequency is 80 Hz. Further, the thresholdvalue TH1 in the present invention is −15 dB from the peak power in theCIR.

FIG. 19 illustrates a delay spread vs. Eb/N0 characteristic thatsatisfies BER=10⁻³. In FIG. 19, the curve A is the characteristic in acase where processing for suppressing background noise is not executed,the curve B is the characteristic in a case where the transmission pathis known (Perfect), the curve C is the characteristic when frequencyaveraging is has been applied to five adjacent subcarriers in the priorart, the curve D is the characteristic according to the presentinvention, in which the propagation-path response vector({overscore (h)}_(t))has been estimated according to Equation (23), and the curve E is thecharacteristic according to the present invention when thepropagation-path response vector has been estimated according to theEquation (20).

It will be understood from the characteristic of the prior art that adelay spread for which the algorithm of the prior art is optimizedexists. Under the present conditions, a delay spread of 0.111 us is theoptimum value and the difference relative to the known propagation pathcharacteristic B (Perfect) is about 0.3 dB. However, the characteristicworsens with distance from the optimum value and is degraded to about4.5 dB at a delay spread of 0.667 us. On the other hand, thecharacteristic D of the present invention is such that under conditionswhere the delay spread is small, degradation from the known propagationpath characteristic B occurs owing to the influence of the adjacentpaths and becomes about 0.6 dB. However, degradation becomes 0.2 dBunder conditions where the delay spread is large. Further, thecharacteristic E of the present invention is such that the difference isalways about 0.1 dB, regardless of the delay spread. Thus it isconfirmed that background noise can be suppressed ideally.

Thus, in accordance with the present invention applied to communicationusing OFDM-based modulation in which subcarriers not used in datatransmission exist, even if a delayed wave that exceeds a guard intervalis generated, it is possible to obtain a propagation path estimationvalue in which background noise is suppressed to a level equivalent tothat in a case where the propagation path is known.

(E) Fourth Embodiment

In the above embodiment, S is defined as the cyclic sinc function matrixdetermined by the relation between the number Nc of propagationsubcarriers and the number N of IFFT points used in OFDM modulation,that is, the number N of FFT points used in demodulation; in a fourthembodiment, S is defined as the cyclic time response function matrixdetermined by the relation between the above N_(C) and M×N, which is anintegral multiple of the number N of FFT points used in demodulation. Inother words, the fourth embodiment is an extension of the first andsecond embodiments, with the case of M=1 corresponding to the first andsecond embodiments. In the first and second embodiments, a limitedrepresentation in terms of the sinc function is employed; this is a timeresponse function in which the positions of subcarriers which do notpropagate are placed at both edges. When there is no propagation atsubcarrier positions other than the edges, the time response vector ofthe matrix S takes on a different form. Even if the form of a timeresponse function is not itself known, all the discrete time responsevalues of the time response function are obtained through actual IFFT,so that the matrix can be created by the same method used for matrix Screation explained in the first embodiment, and so remains a knownquantity.

FIG. 20 is a block diagram of the OFDM communication system of thefourth embodiment. The system configuration is equivalent to that ofFIG. 14, but a portion of the configuration has been omitted, and thepositions of data input to the IFFT unit 14 and data output positionsfrom the FFT unit are clearly indicated. Also, the positions of the P/Sconverter 15 and guard-interval insertion unit 16, as well as thepositions of the guard-interval removal unit 33 and S/P converter 34,are reversed. The propagation path estimation unit 71 in the fourthembodiment has the configuration shown in FIG. 23, described below.

FIG. 21 is an OFDM frame format example; pilot OFDM symbols P,comprising a plurality of pilot symbols, are time-multiplexed with thedata OFDM symbols D₁ to D_(m). A guard interval GI is inserted at thebeginning of each OFDM symbol. The placement of pilot symbols is notlimited to that of FIG. 21, and placement in arbitrary positions ispossible. Also, pilot signals may be placed discretely in the twodimensions of frequency and time.

FIG. 22 explains the relation between the number N of IFFT and FFTpoints and the number Nc of subcarriers used in actual data propagation,and the relation between Nc data items X₀ to X_(NC/2-1) and X_(N-NC/2)to X_(N-1) and IFFT data input terminals. As shown in (A) of FIG. 22, ofthe N subcarriers, it is assumed that data propagates on Nc subcarriers,and moreover that there is no data transmission on the (N-Nc)/2subcarriers on the two sides of the N subcarriers. In this case, thedata propagated on the N_(C) subcarriers is divided into two from thecenter, and as shown in (B) of FIG. 22, is input to N_(C)/2 terminals onboth sides of the N-point IFFT unit 14, inverted with respect to thefrequency axis. That is, no data is input to the central (N-Nc)terminals of the IFFT unit 14. The (N-Nc) non-transmitting subcarriersmay be placed in any manner. The positions of non-transmitting carrierschange only the form of the time response value given by equation (36)described below, but have no effect on this invention. T_(S) is thesampling rate and T_(G) is the guard interval (GI) time; the OFDM symboltime in an OFDM communication system using FFT with N points isT=N·T _(S) +T _(G).

(a) Propagation Path Estimation Unit of the Fourth Embodiment

FIG. 23 shows the block configuration of the propagation path estimationunit 71 in the fourth embodiment. The N-point Fourier transform (FFT)unit 81 performs N-point Fourier transform processing of receivedtime-domain pilot signals to convert to the frequency domain, andoutputs the results from N_(C)/2 terminals on both sides among the Noutput terminals 0 to N-1, as shown in FIG. 24. Upon receiving a pilotsignal which is a known quantity on the receiving side, the impulseresponse measurement unit (zero-forcing channel estimation unit) 82 usesthe FFT output signal to measure the impulse response h_(ZF)(0) toh_(ZF)(N-1) in the propagation path frequency domain due to zeroforcing, and the upsampling unit (oversampling unit) 83 performs M-foldoversampling (where M is an integer greater than or equal to 1, forexample 2) of the observed impulse response h_(zf()0) to h_(ZF)(N-1), asshown in FIG. 25. The M×N-point inverse Fourier transform (IFFT) unit 84converts the M-fold oversampled impulse response to the time domain, andthe valid impulse judgment unit 85 selects impulse responses equal to orgreater than a predetermined threshold value TH from among thetime-domain impulse responses, and generates and outputs an impulseresponse vector, while also outputting a set vector {overscore (l)},similarly to the first embodiment.

The time response vector generation unit 86 generates a time responsevector using time response functions (in the case of the firstembodiment, sinc functions), based on an integral multiple M×N of thenumber N of IFFT points used in OFDM modulation and on the number N_(C)of subcarriers used in actual propagation, the inverse matrix creationunit 87 creates a time response function vector S using the timeresponse vector, the weight generation unit 88 uses the set vector and m(1≦m≦MN) to create a degenerate matrix S′ which reduces the matrix A ofequation (19a) or the time response function matrix S, determines theinverse matrix of the degenerate matrix S′, and outputs the result asthe weight matrix X.

A multiplication unit (propagation path time response estimation unit)89 multiplies the above impulse response vector by the wait matrix X toestimate the propagation path time response. The zero padding unit 90inserts zeros at sampling points at which time response was not obtainedamong the M×N oversampling points, as shown in FIG. 26; the M×N-pointFourier transform (FFT) unit 91 converts the estimated propagation pathtime response into the frequency domain; and the downsampling unit 92,as shown in FIG. 25, performs M-fold downsampling of thefrequency-domain M×N propagation path time responses to obtain Npropagation path responses in the frequency domain, and inputs theresults to the propagation path compensation unit 42 (see FIG. 20).

In the propagation path estimation unit 71, each of the processing unitsperforms processing in either the frequency domain or in the timedomain. In the block configuration, except for the fact that one of thesignals input to the time response vector generation unit 86 is changedfrom N to an M×N, the central inverse matrix operation/estimation block(performing processing in the time domain) is the same as in the firstembodiment. However, in this preprocessing M-fold oversampling isperformed, so that the size of the matrix S in the first embodiment ischanged from N×N to MN×MN. When M=1, the embodiment is the same as thefirst embodiment, and the preceding and following frequency-domainupsampling/downsampling processing units are omitted.

(b) Explanation of Operation of Propagation Path Estimation Unit UsingEquations

When the channel impulse response (CIR) over L propagation paths in amultipath environment is $\begin{matrix}{{g(t)} = {\sum\limits_{l = 0}^{L - 1}\quad{\alpha_{l}{\delta\left( {t - \tau_{l}} \right)}}}} & (26)\end{matrix}$

Here, α₁ and τ₁ represent the complex amplitude and path position of thelth path. At this time, the maximum-delay path is given byτ_(max)=τL-1In this example, the CIR is assumed to satisfy0≦τmax≦τ_(G)

Even in cases where τ_(max)>τG, there is only a difference in therepresentation of equation (27), and there is no change in the essenceof this invention.

Under the above CIR conditions, the N-dimensional received signal vectorin the frequency domain after passing through the propagation paths,y=[y ₀ . . . y _(NC/2-1) y _(NC/2) . . . Y_(N−NC/2−1) y_(N-NC/2) . . . y_(N-1)]^(T)can be represented as follows.y=FFT _(N)(IFFT _(N)(x)

g+{tilde over (w)})  (27)

Here,

is the cyclic convolution operator, and FFT_(N) and IFFT_(N)respectively represent the N-point FFT and IFFT operations, defined asfollows. $\begin{matrix}{{{IFFT}_{N}(x)} = {\frac{1}{N}{\sum\limits_{n = 0}^{N - 1}\quad{{x(n)}{\mathbb{e}}^{j\quad 2\quad\pi\quad{{kn}/N}}}}}} & (28) \\{{{FFT}_{N}(x)} = {\sum\limits_{k = 0}^{N - 1}\quad{{x(k)}{\mathbb{e}}^{{- j}\quad 2\quad\pi\quad{{kn}/N}}}}} & (29)\end{matrix}$

x, g and w are respectively the N-dimensional transmission signalvector, propagation path response vector, and noise vector, given byx=[x ₀ . . . x _(N) _(c) _(/2−1)0 . . . 0x_(N-N) _(c) _(/2) . . .x_(N-1)]^(T)g=[g ₀ g ₁ . . . g _(N-1)]^(T){tilde over (w)}=[{tilde over (w)} ₀ {tilde over (w)} ₁ . . . {tildeover (w)} _(N−1)]^(T)The right-hand side of equation (27) can be modified as follows.y=XF _(N) g+w  (30)

Here X is the diagonal matrix:X=diag(x ₀ . . . X _(N) _(c) _(/2-1)0 . . . 0 x _(N-N) _(c) _(/2) . . .x_(N-1))  (31)

F_(N) is an N×N-dimensional FFT matrix, and the element in the kth rowand nth column is represented by[F _(N)]_(k,n) =e ^(−j2πkn/N) 0≦k,n≦N−1

Further,w=FFT _(N)({tilde over (w)})

From equation (30), using the zero forcing (Least Squares) method, whichis a propagation path estimation method of the prior art, knowntransmission symbols are used to estimate the propagation path frequencyresponse as follows.ĥ _(ZF) =X ⁻¹ y=X ⁻¹(XF _(N) g+w)=F _(N) g+{overscore (W)}  (32)

Here,{overscore (w)}=X⁻¹w

A feature of this invention is that interference components aresuppressed in the time domain.

First, the propagation path frequency response value {hacek over(ĥ_(ZF), estimated using the zero forcing method, is subjected to thefollowing M-fold oversampling. $\begin{matrix}{{\overset{\hat{\Cup}}{h}}_{ZF} = \left\{ \begin{matrix}{{\hat{h}}_{ZF}(n)} & {0 \leq n < {N_{c}/2}} \\0 & {{N_{c}/2} \leq n < {{MN} - {N_{c}/2}}} \\{{\hat{h}}_{ZF}\left( {n - {\left( {M - 1} \right)N}} \right)} & {{{MN} - {N_{c}/2}} \leq n < {MN}}\end{matrix} \right.} & (33)\end{matrix}$

Here N_(C) is the number of subcarriers used in transmission.

Next, the frequency response, oversampled in the frequency domain, isconverted into the time domain by M×N-point IFFT. This converted signal{hacek over (ĝ_(ZF) can be expressed as follows.{hacek over (ĝ _(ZF) =IFFT _(MN)({hacek over (ĥ _(ZF))=S{hacek over(g)}+{overscore ({hacek over (w)})}  (34)

Here {hacek over (g)} is the true time response vector at an M-foldoversampled point, and S is a matrix according to the time responsefunctions, taking into account non-transmitting subcarriers, and definedby the following equation. $\begin{matrix}{S = \begin{bmatrix}{s(0)} & {s(1)} & \cdots & {s\left( {{MN}/2} \right)} & \cdots & {s(1)} \\{s(1)} & ⋰ & ⋰ & \quad & ⋰ & \vdots \\\vdots & ⋰ & ⋰ & ⋰ & \quad & {s\left( {{MN}/2} \right)} \\{s\left( {{MN}/2} \right)} & \quad & ⋰ & ⋰ & ⋰ & \vdots \\\vdots & ⋰ & \quad & ⋰ & ⋰ & {s(1)} \\{s(1)} & \cdots & {s\left( {{MN}/2} \right)} & \cdots & {s(1)} & {s(0)}\end{bmatrix}} & (35)\end{matrix}$

Here, if both edges are assumed to be non-transmitting, s(n) is definedby the following equation. In this example, a sinc function sample pointrepresentation is used, but a cyclic matrix S of time response vectorsis uniquely determined corresponding to an arbitrary pilot arrangement.That is, it is possible to prepare the matrix S as known information onthe receiving side. $\begin{matrix}{{s(n)} = {\frac{1}{MN} \cdot \frac{\sin\left( {\pi\quad N_{c}\frac{n}{MN}} \right)}{\pi\quad N_{c}\frac{n}{MN}} \cdot N_{c}}} & (36)\end{matrix}$

The inverse matrix of the matrix S is found from equation (35), and bymultiplying from the left of equation (34), the estimated value of g canbe obtained.{hacek over (ĝ _(IM) =S ⁻¹ {hacek over (ĝ _(ZF) =S ⁻¹(S{hacek over(g)}+{overscore ({hacek over (w)})})={hacek over (g)}+S ⁻¹ {overscore({hacek over (w)})}  (37)

However, an extremely large amount of computation is necessary to findthe inverse matrix of S. Hence the degenerate matrix equation ofequation (35) is generated in order to enable suppression ofinterference components represented by S⁻¹{overscore ({hacek over (w)})}and to reduce the amount of computation. To this end, it is noted thatin a matrix conforming to the time response function represented byequation (35) such as the sinc function, the power drops off rapidly inmoving away from the diagonal components.

The correlation between {hacek over (g)} and {hacek over (ĝ_(ZF) ishigh. This is probably because {hacek over (ĝ_(ZF) is the result offiltering {hacek over (g)} by S. Here, among the components of {hacekover (ĝ_(ZF), the components below the predetermined threshold value THare removed, to produce a degenerate matrix as indicated below. First,in equation (34), if the kth sample {hacek over (ĝ_(ZF)(k) is assumed tobe a component below TH, then this sample, and the right-hand sidecomponents {hacek over (g)}(k) and {overscore ({hacek over (w)})}(k),corresponding to this, are deleted.

Next, the submatrix S′ is generated by deleting rows and columns in Scorresponding to the deleted samples. This processing is mathematicallycalled “creating a submatrix, a simple example of which is given below.If M=1 and N=4, then equation (34) is expressed as the followingequation $\begin{bmatrix}{{\overset{\hat{\Cup}}{g}}_{ZF}(0)} \\{{\overset{\hat{\Cup}}{g}}_{ZF}(1)} \\{{\overset{\hat{\Cup}}{g}}_{ZF}(2)} \\{{\overset{\hat{\Cup}}{g}}_{ZF}(3)}\end{bmatrix} = {{\begin{bmatrix}{s(0)} & {s(1)} & {s(2)} & {s(1)} \\{s(1)} & {s(0)} & {s(1)} & {s(2)} \\{s(2)} & {s(1)} & {s(0)} & {s(1)} \\{s(1)} & {s(2)} & {s(1)} & {s(0)}\end{bmatrix} \cdot \begin{bmatrix}{\overset{\Cup}{g}(0)} \\{\overset{\Cup}{g}(1)} \\{\overset{\Cup}{g}(2)} \\{\overset{\Cup}{g}(3)}\end{bmatrix}} + \begin{bmatrix}{\overset{\overset{\Cup}{\_}}{w}(0)} \\{\overset{\overset{\Cup}{\_}}{w}(1)} \\{\overset{\overset{\Cup}{\_}}{w}(2)} \\{\overset{\overset{\Cup}{\_}}{w}(3)}\end{bmatrix}}$

Here, it is assumed that samples estimated using ZF (zero forcing) belowthe threshold TH provided by the system are{hacek over (ĝ_(ZF)(2) and {hacek over (ĝZF(3).

First, the corresponding right-hand side,{hacek over (ĝ(2) and {hacek over (ĝ(3), {overscore ({hacek over(w)})}(2) and {overscore ({hacek over (w)})}(3)are deleted. Next, the rows and columns in S corresponding to thesedeleted samples are deleted, so that the above matrix equation producesthe degenerate matrix $\begin{bmatrix}{{\overset{\hat{\Cup}}{g}}_{ZF}(0)} \\{{\overset{\hat{\Cup}}{g}}_{ZF}(1)}\end{bmatrix} = {{\begin{bmatrix}{s(0)} & {s(1)} \\{s(1)} & {s(0)}\end{bmatrix} \cdot \begin{bmatrix}{\overset{\Cup}{g}(0)} \\{{\overset{\Cup}{g}}_{t}(1)}\end{bmatrix}} + \begin{bmatrix}{\overset{\overset{\Cup}{\_}}{w}(0)} \\{\overset{\overset{\Cup}{\_}}{w}(1)}\end{bmatrix}}$

Here, the submatrix is $S^{\prime} = \begin{bmatrix}{s(0)} & {s(1)} \\{s(1)} & {s(0)}\end{bmatrix}$

The above degenerate matrix equation is the same as equation (22), andthis example is in essence the same as the example presented for thefirst embodiment.

If the matrix equation resulting from degeneracy due to submatrixcreation is defined as{hacek over (ĝ′ _(ZF) =S′{hacek over (g)}′+{overscore ({hacek over(w)})}′  (37)then by determining the inverse matrix of S′, it is possible tocalculate {hacek over (g)}′. The size of S′ is extremely small comparedwith S, and the following equation{hacek over (ĝ′ _(DIM) =S′ ⁻¹ {hacek over (ĝ′ _(ZF)(38)obtains. Moreover, by substituting 0 (zero padding) for samples otherthan the obtained {hacek over (ĝ′_(DIM), it is possible to suppressinterference components.

If the time response value of propagation paths after zero-substitutionis completed is {circumflex over ({hacek over (g)})}_(DIM), then thefrequency response value can be represented by the equation{hacek over (ĥ _(DIM) =FFT _(MN)({hacek over (ĝ _(DIM))(39)

Hence by performing downsampling according to equation (40), thefrequency response value of propagation paths with interferencecomponents suppressed can be obtained. $\begin{matrix}{{\hat{h}}_{DIM} = \left\{ \begin{matrix}{{\overset{\hat{\Cup}}{h}}_{DIM}(n)} & {0 \leq n < {N_{c}/2}} \\{{\overset{\hat{\Cup}}{h}}_{DIM}\left( {n + {\left( {M - 1} \right)N}} \right)} & {{N - {N_{c}/2}} \leq n < N}\end{matrix} \right.} & (40)\end{matrix}$

(c) Advantageous Results of the Fourth Embodiment

By means of the fourth embodiment, characteristic degradation isprevented even in more realistic propagation path environments. That is,multipath propagation paths, occurring as physical phenomena, occur forcontinuous signals, that is, for analog signals. The positions of eachof the paths in such a case are not necessarily the sampling positionsmeasured by the system, that is, the sampling intervals, and it isanticipated that deviation from these positions will occur frequently inreal environments. It should be noted that a channel mapped to asampling interval is called a “sample-spaced channel”.

In the path model used in simulations, an L-path exponential functionmode is defined to simulate a real environment. As shown in FIG. 27,each of the L paths is positioned at τ_(l)(0≦l≦L−1), and Ith pathattenuates as (1×ΔP)dB in comparison with an advance wave. Further, thepath intervals are defined by Δτ=τ_(i)−τ_(i−1)

Here, a channel for which there is not even one path at the sampledposition is called a “non-sample-spaced channel”, and in equation (26),cases obtain in which there exist one or more paths for whichτ_(l)/T_(S) is not an integer. In this channel model, it is known thatcharacteristics are degraded for methods such as that of this inventionin which channel estimation is performed in the time domain.

Here, two channel models are studied, and the advantageous results ofthe fourth embodiment are clarified.

-   1) Channel A: τ₀=0, Δτ=18 (samples) (“sample-spacedchannel”)-   2) Channel B: τ₀=T_(S)/2, Δτ=18 (samples) (“non-sample-spaced    channel”)

Here, Channel B is a real environment in which all paths doe not existat the sampled position, and, for the fourth embodiment, can define theworse-case conditions. The simulation conditions appear in Table 1.Here, the threshold value TH is taken to be −16 dB from a sample havingthe maximum power. TABLE 1 Simulation parameters Carrier frequency    5GH_(Z) Sampling frequency 78.34 MH_(Z) Number of subcarriers   896IFFI/FFT Point number  1024 Subcarrier interval  76.5 kH_(Z) Symbolinterval 15.63 us GI length  2.55 us(200 samples) Frame length  0.5 msModulation method   16 QAM Error-correction code Turbocode (R = ½, K =4) Max-Log-MAP (iterations = 8) decoding Reception diversity YesPropagation path model L = 12 path exponential attenuation model (FIG.4) ΔP = 1 [dB] Rayleigh fading (fd = 480 H_(Z))

FIG. 28 (Channel A) and FIG. 29 (Channel B) show the Eb/No versus BERcharacteristics when using 16QAM. The characteristics of the firstembodiment with M=1 and the fourth embodiment with M=2 are respectivelydisplayed. From FIG. 28, in Channel A for a BER of 10⁻³ with propagationpath known (perfect channel-state information, or perfect CSI),degradation of 0.1 dB for the first embodiment with M=1 and of 0.2 dBfor the second embodiment are observed. However, as is clear from FIG.29, for Channel B which is more nearly like a real environment,characteristic degradation does not occur for perfect CSI in the fourthembodiment. In the case of the first embodiment with M=1, it is seenthat considerable characteristic degradation (0.9 dB at a BER of 10⁻³)occurs. Further, FIG. 30 (Channel A) and FIG. 31 (Channel B) show theEb/No versus BER characteristics when using 64QAM. The characteristicsof the first embodiment with M=1 and the fourth embodiment with M=2 arerespectively displayed. Using 64QAM, with lower noise tolerance than16QAM, the characteristic degradation for perfect CSI is considerableeven for Channel A, as shown in FIG. 30. That is, for BER=10⁻³,degradation of 0.3 dB for perfect CSI with M=1 in the first embodiment,and of 0.4 dB with M=2 in the second embodiment, are observed. However,for Channel B which is more nearly like a real environment, as is clearfrom FIG. 31, there is no change in the characteristic difference forperfect CSI in the second embodiment with M=2. But in the case of thefirst embodiment with M=1, it is seen that considerable characteristicdegradation occurs (3.7 dB at BER=10⁻³). From the above simulations, bymeans of the fourth embodiment, noise interference can be suppressed andcharacteristics can be improved even in channel estimation in realenvironments in which no paths exist at sampled positions.

As many apparently widely different embodiments of the present inventioncan be made without departing from the spirit and scope thereof, it isto be understood that the invention is not limited to the specificembodiments thereof except as defined in the appended claims.

Additional Note

-   1. A propagation path estimation method of a receiver in an OFDM    (Orthogonal Frequency Division Multiplexing) communication system    for performing communication by OFDM, comprising the steps of:    -   estimating a group of impulse responses (CIRs) of a propagation        paths;    -   selecting impulse responses, which are greater than a        predetermined threshold value, from the impulse-response group;    -   substituting a predetermined value for samples other than a        prescribed number of samples bracketing a maximum peak in each        impulse response selected; and    -   estimating the propagation paths using the impulse responses        obtained by substitution.-   2. The method according to item 1, wherein said step of estimating    the group of impulse responses (CIRs) includes multiplying a    receive-signal vector by a conjugate transposed matrix of known    pilot symbols and estimating, on a per propagation-path basis,    propagation-path impulse responses (CIRs) comprising a plurality of    samples of a time series.-   3. The method according to item 1, wherein said step of estimating    the propagation paths includes the steps of:    -   obtaining an inverse matrix of a sinc-function matrix (S matrix)        decided based upon a number N of points of an IFFT used in OFDM        modulation and number Nc of subcarriers used in actual        transmission;    -   multiplying a CIR estimation vector composed of the CIRs by the        inverse matrix to thereby calculate a propagation-path response        vector; and    -   estimating a characteristic of the propagation path from the        propagation-path response vector.-   4. The method according to item 3, wherein the propagation path is    estimated by adopting said prescribed values for those elements of    the propagation-path response vector obtained by calculation that    are less than a threshold value.-   5. The method according to item 1, wherein the prescribed number of    samples is decided based upon a number N of points of an IFFT used    in OFDM modulation and number Nc of subcarriers used in actual    transmission.-   6. A propagation path estimation method of a receiver in an OFDM    (Orthogonal Frequency Division Multiplexing) communication system    for performing communication by OFDM, comprising the steps of:    -   estimating a group of impulse responses (CIRs) of propagation        paths;    -   selecting propagation-path impulse responses (CIRs), which are        greater than a predetermined threshold value, from the        propagation-path impulse-response group;    -   generating a matrix expression using a CIR estimation vector        ({circumflex over ({overscore (h)})}_(CIR))        that includes the selected CIRs as elements, a matrix S, which        is decided based upon number N of points of an IFFT used in OFDM        modulation and number Nc of subcarriers used in actual        transmission, and a propagation-path response vector        ({overscore (h)}_(t))    -   ; and    -   obtaining the propagation-path response vector by solving this        matrix expression.-   7. The method according to item 6, wherein said step of estimating    the group of propagation-path impulse responses (CIRs) includes    multiplying a receive-signal vector by a conjugate transposed matrix    of known pilot symbols and estimating, on a per-propagation-path    basis, a CIR estimation vector    ({circumflex over ({overscore (h)})}_(CIR))    comprising a plurality of samples of a time series.-   8. The method according to item 6, wherein the matrix S is a    sinc-function matrix decided based upon the number N of points of    IFFT and number Nc of subcarriers.-   9. The method according to item 6, wherein said step of generating    the matrix expression includes generating the matrix expression as    {circumflex over ({overscore (h)})} _(CIR) =S·{overscore (h)} _(t)    +P _(t) *·{overscore (w)}    (where P_(t)* is a conjugate transposed matrix of known pilot    symbols) taking a noise power vector {right arrow over (w)} into    account, and obtaining the propagation-path response vector from    this matrix expression.-   10. The method according to item 6, wherein if propagation-path    impulse responses (CIR) greater than the threshold value can be    divided into a plurality of blocks, the matrix expression is    generated block by block to thereby obtain the propagation-path    response vector.-   11. The method according to item 9, wherein if there is even a    single propagation-path impulse response (CIR) that belongs to the    blocks, then the propagation-path response vector is found by    multiplying the CIR estimation vector by a fixed value, which is    decided based upon the number N of points and number Nc of    subcarriers, without generating a matrix expression.-   12. The method according to item 6, wherein said step of obtaining    the propagation-path response vector includes the steps of:    -   obtaining an inverse matrix of the matrix S, which is a        sinc-function decided based upon the number N of points of the        IFFT and number Nc of subcarriers; and    -   multiplying the CIR estimation vector by the inverse matrix to        thereby obtain the propagation-path response vector.-   13. The method according to item 6, wherein said step of obtaining    the propagation-path response vector includes the steps of:    -   obtaining a matrix that is in accordance with standard of an        MMSE (Minimum Mean Square Error) using the matrix S, which is a        sinc-function matrix decided based upon the number N of points        of the IFFT and number Nc of subcarriers, and variance of noise;        and    -   multiplying the CIR estimation vector by the matrix to thereby        obtain the propagation-path response vector.-   14. The method according to item 12 or 13, wherein the propagation    path is estimated by adopting prescribed values for those elements    of the propagation-path response vector obtained by calculation that    are less than a threshold value.-   15. The method according to item 6 or 14, wherein the threshold    value is made power that is lower than a maximum peak value of the    CIR by a value decided based upon the number N of points of the IFFT    used in OFDM modulation and the number Nc of subcarriers used in    actual transmission, or is made power that is greater than estimated    background noise power by a predetermined value.-   16. A propagation path estimation apparatus of a receiver in an OFDM    (Orthogonal Frequency Division Multiplexing) communication system    for performing communication by OFDM, comprising:    -   a CIR estimation unit for estimating a group of impulse        responses (CIRs) of propagation paths;    -   a valid-impulse discriminator for selecting impulse responses,        which are greater than a predetermined threshold value, from the        impulse-response group and substituting predetermined value for        samples other than a prescribed number of samples bracketing a        maximum peak in each impulse response selected; and    -   a propagation path estimation unit for estimating the        propagation path using the valid impulse responses.-   17. The apparatus according to item 16, wherein said CIR estimation    unit multiplies a receive-signal vector by a conjugate transposed    matrix of a known symbol and estimates, on a per-propagation-path    basis, propagation-path impulse responses (CIRs) comprising a    plurality of samples of a time series.-   18. The apparatus according to item 16, wherein said propagation    path estimation unit includes:    -   means for acquiring a sinc-function matrix (S matrix) decided        based upon a number N of points of an IFFT used in OFDM        modulation and number Nc of subcarriers used in actual        transmission, and an inverse matrix of this matrix;    -   means for multiplying a CIR estimation vector composed of the        CIRs by the inverse matrix to thereby calculate a        propagation-path response vector.-   19. The apparatus according to item 18, wherein said propagation    path estimation unit includes means for estimating the propagation    path by adopting said prescribed values for those elements of the    propagation-path response vector obtained by calculation that are    less than a threshold value.-   20. A propagation path estimation apparatus of a receiver in an OFDM    (Orthogonal Frequency Division Multiplexing) communication system    for performing communication by OFDM, comprising:    -   a CIR estimation unit for estimating a group of impulse        responses (CIRs) of propagation paths;    -   a valid-impulse discriminator for selecting propagation-path        impulse responses (CIRs), which are greater than a predetermined        threshold value, from the propagation-path impulse-response        group; and    -   a propagation path estimation unit for generating a matrix        expression using a CIR estimation vector        ({circumflex over ({overscore (h)})}_(CIR))        that includes the selected CIRs as elements, a matrix S, which        is decided based upon number N of points of an IFFT used in OFDM        modulation and number Nc of subcarriers used in actual        transmission, and a propagation-path response vector        ({overscore (h)}_(t))        , obtaining the propagation-path response vector by solving this        matrix expression and estimating the propagation path using this        response vector.-   21. The apparatus according to item 20, wherein said CIR estimation    unit includes means for multiplying a receive-signal vector by a    conjugate transposed matrix of known pilot symbols and estimating,    on a per-propagation-path basis, a CIR estimation vector    ({circumflex over ({overscore (h)})}_(CIR))    comprising a plurality of samples of a time series.-   22. The apparatus according to item 20, wherein said    propagation-path estimation unit includes:    -   means for calculating an inverse matrix of the matrix S, which        is a sinc-function decided based upon the number N of points of        the IFFT and number Nc of subcarriers; and    -   means for multiplying the CIR estimation vector by the inverse        matrix to thereby obtain the propagation-path response vector.-   23. The apparatus according to item 20, wherein said    propagation-path estimation unit includes:    -   means for obtaining a matrix that is in accordance with standard        of an MMSE (Minimum Mean Square Error) using the matrix S, which        is a sinc-function matrix decided based upon the number N of        points of the IFFT and number Nc of subcarriers, and variance of        noise; and    -   means for multiplying the CIR estimation vector by the matrix to        thereby obtain the propagation-path response vector.-   24. The apparatus according to item 22 or 23, wherein said    propagation-path estimation unit further includes means for    estimating the propagation path by adopting prescribed values for    those elements of the propagation-path response vector obtained by    calculation that are less than a threshold value.-   25. A path searcher of a RAKE receiver in a CDMA communication    system, comprising:    -   a CIR estimation unit for estimating a group of impulse        responses (CIRs) of propagation paths;    -   a valid-impulse discriminator for selecting propagation-path        impulse responses (CIRs), which are greater than a predetermined        threshold value, from the propagation-path impulse-response        group; and    -   a propagation path estimation unit for estimating the        propagation path using the CIRs;    -   wherein said propagation path estimation unit generates a matrix        expression using a CIR estimation vector        ({circumflex over ({overscore (h)})}_(CIR))        that includes the selected CIRs as elements, a matrix S, which        is decided by impulse responses of a low-pass filter, and a        propagation-path response vector        ({overscore (h)}_(t))        and searching for a propagation path by obtaining the        propagation-path response vector by solving this matrix        expression.-   26. A propagation path estimation method of a receiver in an OFDM    communication system which communicates using an orthogonal    frequency division multiplexing (OFDM), comprising the steps of:-   estimating impulse responses of propagation paths in the frequency    domain;-   subjecting the estimated impulse responses to M-fold oversampling    (where M is an integer greater than or equal to 1);-   converting the M-fold oversampled impulse responses into the time    domain;-   selecting, among the time-domain impulse responses, impulse    responses equal to or greater than a predetermined threshold value;-   replacing everything other than a prescribed number of samples    before and after the maximum peak in the selected impulse responses    with a prescribed value;-   estimating propagation path time response using impulse responses    obtained from said replacement; and,-   after converting estimated time responses into the frequency domain,    performing M-fold downsampling and estimating the propagation path.-   27. An OFDM propagation path estimation method of receiver in an    OFDM communication system which communicates using an orthogonal    frequency division multiplexing (OFDM) comprising steps of:-   estimating impulse responses of propagation paths in the frequency    domain;-   subjecting the estimated impulse responses to M-fold oversampling    (where M is an integer greater than or equal to 1);-   converting the M-fold oversampled impulse responses into the time    domain;-   selecting, among the time-domain impulse responses, impulse    responses equal to or greater than a predetermined threshold value,    and generating an impulse response vector; creating a time response    function matrix according to time response functions, based on an    integral multiple M□N of the number N of IFFT points used in OFDM    conversion and on the number NC of subcarriers used actually in    propagation, and multiplying the inverse matrix thereof by said    impulse response vector to estimate the propagation path time    response; and,-   after converting the estimated time response into the frequency    domain, performing M-fold downsampling and estimating the    propagation path.-   28. The OFDM propagation path estimation method according to item    27, wherein replacing everything other than a prescribed number of    samples before and after the maximum peak in said time response    function by a prescribed value so as to create said time response    function matrix.-   29. The OFDM propagation path estimation method according to item 27    or item 28, wherein time-domain pilot reception signals are    subjected to N-point Fourier transform processing and are converted    into the frequency domain, and the frequency-domain signals are used    to estimate the propagation path impulse response in the frequency    domain.-   30. The OFDM propagation path estimation method according to item 26    or item 27, wherein said M-fold oversampled impulse response is    converted into a time-domain impulse response by M×N-point inverse    Fourier transfer processing, and said propagation path time response    is converted into a frequency-domain propagation path time response    by M×N-point Fourier transform processing.-   31. A propagation path estimation apparatus of a receiver in an OFDM    communication system which communicates using an orthogonal    frequency division multiplexing (OFDM), comprising:-   an impulse response estimation unit, which estimates impulse    responses in the frequency domain of propagation paths;-   an oversampling unit, which performs M-fold oversampling (where M is    an integer greater than or equal to 1) of the estimated impulse    responses;-   an inverse Fourier transform unit, which converts the M-fold    oversampled impulse responses into the time domain;-   a valid impulse judgment unit, which selects, from among the    time-domain impulse responses, impulse responses-   greater than or equal to a predetermined threshold value; an    estimation unit, which replaces everything other than a prescribed    number of samples before and after the maximum peak in the selected    impulse responses by a prescribed value;-   a Fourier transform unit, which converts the estimated propagation    path time responses into the frequency domain; and,-   a propagation path estimation unit, which performs M-fold    downsampling of the time response in the frequency domain and    estimates the propagation path.-   32. A propagation path estimation apparatus of a receiver in an OFDM    communication system which communicates using an orthogonal    frequency division multiplexing (OFDM), comprising:-   an impulse response estimation unit, which estimates impulse    responses in the frequency domain of propagation paths;-   an oversampling unit, which performs M-fold oversampling (where M is    an integer greater than or equal to 1) of the estimated impulse    responses;-   an inverse Fourier transform unit, which converts the M-fold    oversampled impulse responses into the time domain; a valid impulse    judgment unit, which selects, from among the time-domain impulse    responses, impulse responses greater than or equal to a    predetermined threshold value, and generates an impulse response    vector;-   a propagation path time impulse estimation unit, which creates a    time response function matrix using a time response function based    on an integral multiple M□N of the number N of IFFT points used in    OFDM modulation and on the number Nc of subcarriers used in actual    propagation, and which estimates the propagation path time response    by multiplying the inverse matrix thereof by said impulse response    vector;-   a Fourier transform unit, which converts the estimated propagation    path time response into the frequency domain; and,-   means for performing M-fold downsampling of the propagation path    time response in the frequency domain and for estimating the    propagation path.-   33. The propagation path estimation apparatus according to item 32,    wherein said propagation path time response estimation unit creates    said time response function matrix such that everything other than a    prescribed number of samples before and after the maximum peak in    said selected impulse response is replaced with a prescribed value.-   34. The propagation path estimation apparatus according to item 31    or item 32, further comprising a Fourier transform unit which    subjects time-domain pilot reception signals to N-point Fourier    transform processing for conversion to the frequency domain, and    wherein said impulse response estimation unit uses the    frequency-domain signals to estimate the propagation path impulse    response of propagation paths in the frequency domain.

1. A propagation path estimation apparatus of a receiver in an OFDM(Orthogonal Frequency Division Multiplexing) communication system forperforming communication by OFDM, comprising: a CIR estimation unit forestimating a group of impulse responses (CIRs) of propagation paths; avalid-impulse discriminator for selecting impulse responses, which aregreater than a predetermined threshold value, from the impulse-responsegroup and substituting predetermined value for samples other than aprescribed number of samples bracketing a maximum peak in each impulseresponse selected; and a propagation path estimation unit for estimatingthe propagation path using the valid impulse responses.
 2. The apparatusaccording to claim 1, wherein said CIR estimation unit multiplies areceive-signal vector by a conjugate transposed matrix of a known symboland estimates, on a per-propagation-path basis, propagation-path impulseresponses (CIRs) comprising a plurality of samples of a time series. 3.The apparatus according to claim 1, wherein said propagation pathestimation unit includes: means for acquiring a sinc-function matrix (Smatrix) decided based upon a number N of points of an IFFT used in OFDMmodulation and number Nc of subcarriers used in actual transmission, andan inverse matrix of this matrix; means for multiplying a CIR estimationvector composed of the CIRs by the inverse matrix to thereby calculate apropagation-path response vector.
 4. The apparatus according to claim 3,wherein said propagation path estimation unit includes means forestimating the propagation path by adopting said prescribed values forthose elements of the propagation-path response vector obtained bycalculation that are less than a threshold value.
 5. A propagation pathestimation apparatus of a receiver in an OFDM (Orthogonal FrequencyDivision Multiplexing) communication system for performing communicationby OFDM, comprising: a CIR estimation unit for estimating a group ofimpulse responses (CIRs) of propagation paths; a valid-impulsediscriminator for selecting propagation-path impulse responses (CIRs),which are greater than a predetermined threshold value, from thepropagation-path impulse-response group; and a propagation pathestimation unit for generating a matrix expression using a CIRestimation vector({circumflex over ({overscore (h)})}_(CIR)) that includes the selectedCIRs as elements, a matrix S, which is decided based upon number N ofpoints of an IFFT used in OFDM modulation and number Nc of subcarriersused in actual transmission, and a propagation-path response vector({overscore (h)}_(t)) , obtaining the propagation-path response vectorby solving this matrix expression and estimating the propagation pathusing this response vector.
 6. The apparatus according to claim 5,wherein said CIR estimation unit includes means for multiplying areceive-signal vector by a conjugate transposed matrix of known pilotsymbols and estimating, on a per-propagation-path basis, a CIRestimation vector({circumflex over ({overscore (h)})}_(CIR)) comprising a plurality ofsamples of a time series.
 7. The apparatus according to claim 5, whereinsaid propagation-path estimation unit includes: means for calculating aninverse matrix of the matrix S, which is a sinc-function decided basedupon the number N of points of the IFFT and number Nc of subcarriers;and means for multiplying the CIR estimation vector by the inversematrix to thereby obtain the propagation-path response vector.
 8. Theapparatus according to claim 5, wherein said propagation-path estimationunit includes: means for obtaining a matrix that is in accordance withstandard of an MMSE (Minimum Mean Square Error) using the matrix S,which is a sinc-function matrix decided based upon the number N ofpoints of the IFFT and number Nc of subcarriers, and variance of noise;and means for multiplying the CIR estimation vector by the matrix tothereby obtain the propagation-path response vector.
 9. The apparatusaccording to claim 7, wherein said propagation-path estimation unitfurther includes means for estimating the propagation path by adoptingprescribed values for those elements of the propagation-path responsevector obtained by calculation that are less than a threshold value. 10.A path searcher of a RAKE receiver in a CDMA communication system,comprising: a CIR estimation unit for estimating a group of impulseresponses (CIRs) of propagation paths; a valid-impulse discriminator forselecting propagation-path impulse responses (CIRs), which are greaterthan a predetermined threshold value, from the propagation-pathimpulse-response group; and a propagation path estimation unit forestimating the propagation path using the CIRS; wherein said propagationpath estimation unit generates a matrix expression using a CIRestimation vector({circumflex over ({overscore (h)})}_(CIR)) that includes the selectedCIRs as elements, a matrix S, which is decided by impulse responses of alow-pass filter, and a propagation-path response vector({overscore (h)}_(t)) and searching for a propagation path by obtainingthe propagation-path response vector by solving this matrix expression.11. A propagation path estimation method of a receiver in an OFDMcommunication system which communicates using an orthogonal frequencydivision multiplexing (OFDM), comprising the steps of: estimatingimpulse responses of propagation paths in the frequency domain;subjecting the estimated impulse responses to M-fold oversampling (whereM is an integer greater than or equal to 1); converting the M-foldoversampled impulse responses into the time domain; selecting, among thetime-domain impulse responses, impulse responses equal to or greaterthan a predetermined threshold value; replacing everything other than aprescribed number of samples before and after the maximum peak in theselected impulse responses with a prescribed value; estimatingpropagation path time response using impulse responses obtained fromsaid replacement; and, after converting estimated time responses intothe frequency domain, performing M-fold downsampling and estimating thepropagation path.
 12. The propagation path estimation apparatusaccording to claim 11, wherein the valid impulse judgment unit,generates an impulse response vector by selecting impulse responsesgreater than or equal to a predetermined threshold value and thepropagation path time impulse estimation unit, creates a time responsefunction matrix using a time response function based on an integralmultiple M×N of the number N of IFFT points used in OFDM modulation andon the number Nc of subcarriers used in actual propagation, andestimates the propagation path time response by multiplying the inversematrix thereof by said impulse response vector.
 13. The propagation pathestimation apparatus according to claim 12, wherein said propagationpath time response estimation unit creates said time response functionmatrix such that everything other than a prescribed number of samplesbefore and after the maximum peak in said selected impulse response isreplaced with a prescribed value.
 14. The propagation path estimationapparatus according to claim 11, further comprising a Fourier transformunit which subjects time-domain pilot reception signals to N-pointFourier transform processing for conversion to the frequency domain, andwherein said impulse response estimation unit uses the frequency-domainsignals to estimate the propagation path impulse response of propagationpaths in the frequency domain.